Method and apparatus for receive beamformer system

ABSTRACT

The present invention includes a fully programmable plurality of multi-channel receivers, each receiver having a digital multi-channel receive processor and a local processor control. Each receive processor includes a first decimator, time delay memory, second decimator, and complex multiplier. The receive beamformer is a computationally efficient system which is programmable to allow processing mode trade-offs among receive frequency, receive spatial range resolution, and number of simultaneous beams received. Each local control receives focusing data from a central control computer and provides final calculation of per-channel dynamic focus delay, phase, apodization, and calibration values for each receiver signal sample. Further, this invention includes a baseband multi-beam processor which has a phase aligner and a baseband filter for making post-beamformation coherent phase adjustments and signal shaping, respectively. The phase aligner maintains scan-line-to-scan-line coherency, such as would be required for coherent image formation. Accordingly, the present system can operate under multiple imaging formats, using a variety of transducers and a variety of modes such as B-mode, M-mode, and color Doppler flow mode.

REFERENCE TO RELATED APPLICATION

This application is a divisional Ser. No. 08/432,615 filed May 2, 1995and now U.S. Pat. No. 5,685,308, continuation in part of Ser. No.08/286,658 filed Aug. 5, 1994, and now abandoned.

REFERENCE TO MICROFICHE APPENDIX

This application includes a microfiche appendix of 195 sheets ofmicrofiche having 19,058 frames. A portion of the disclosure of thispatent document contains material which is subject to copyrightprotection. The copyright owner has no objection to the facsimilereproduction by any one of the patent disclosure, as it appears in thePatent and Trademark Office patent files or records, but otherwisereserves all copyright rights whatsoever.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to:

a. METHOD AND APPARATUS FOR TRANSMIT BEAMFORMER SYSTEM, Cole et al.,Attorney Docket No. 5055-78;

b. METHOD AND APPARATUS FOR FOCUS CONTROL OF TRANSMIT AND RECEIVEBEAMFORMER SYSTEMS, Gee et al., Attorney Docket No. 5055-79;

c. METHOD AND APPARATUS FOR DOPPLER RECEIVE BEAMFORMER SYSTEM, Maslak etal., Attorney Docket No. 5055-80;

d. METHOD AND APPARATUS FOR ADJUSTABLE FREQUENCY SCANNING IN ULTRASOUNDIMAGING, Wright et al., Attorney Docket No. 5055-83;

e. METHOD AND APPARATUS FOR A BASEBAND PROCESSOR OF A RECEIVE BEAMFORMERSYSTEM, Wright et al., Attorney Docket No. 5055-84;

f. METHOD AND APPARATUS FOR BEAMFORMER SYSTEM WITH VARIABLE APERTURE,Cole et al., Attorney Docket No. 5055-85.

The above applications are all commonly assigned with the presentapplication, filed concurrently with the present application, and areincorporated herein by reference in their entirety.

The present application is also related to the following previouslyfiled applications:

a. METHOD AND APPARATUS FOR REAL-TIME, CONCURRENT ADAPTIVE FOCUSING INAN ULTRASOUND BEAMFORMER IMAGING SYSTEM, Wright et al., Ser. No.08/286,528, filed Aug. 5, 1994;

b. METHOD AND APPARATUS FOR A GEOMETRIC ABERRATION TRANSFORM IN ANADAPTIVE FOCUSING ULTRASOUND BEAMFORMER SYSTEM, Wright et al., Ser. No.08/286,664, filed Aug. 5, 1994;

c. METHOD AND APPARATUS FOR COHERENT IMAGE FORMATION, Wright et al.,Ser. No. 08/286,510, filed Aug. 5, 1994.

I. FIELD OF THE INVENTION

This invention relates to coherent imaging systems including, forexample, radar, sonar, seismic, and ultrasound systems, using vibratoryenergy, and in particular, but not limited to, phased array ultrasoundimaging systems for scan formats such as linear, steered linear, sector,circular, Vector®, steered Vector® and other types of scan formats inimaging modes such as, by way of example only, B-mode (gray-scaleimaging mode), F-mode (flow or color Doppler imaging mode), M-mode(motion mode) and D-mode (spectral Doppler mode). Although the inventionwill be discussed with respect to an ultrasound system, the inventioncan be implemented with other types of coherent imaging systems.

II. BACKGROUND OF THE INVENTION

A. Literature:

The open literature, which presents issues relevant to imaging systemsin general, includes the following documents which are incorporatedherein by reference:

1. Dan E. Dudgeon, "Fundamentals of Digital Array Processing,"Proceedings of the IEEE, volume 65, pp. 898-904, June 1977.

2. Dan E. Dudgeon and Russell M. Mersereau, Multidimensional DigitalSignal Processing, Chapter 6, Section 2: "Beamforming," Prentice Hall,1984.

3. William C. Knight, Roger G. Pridham, and Steven M. Kay, "DigitalSignal Processing for Sonar," Proceedings of the IEEE, volume 69, pages1451-1506, November 1981. (Digital beamformers for use in sonardescribed on pages 1465-1471.)

4. Roger G. Pridham and Ronald A. Mucci, "A Novel Approach to DigitalBeamforming," Journal of the Acoustical Society of America, volume 63,pages 425-434, February 1978.

5. Roger G. Pridham and Ronald A. Mucci, "Digital InterpolationBeamforming for Low-Pass and Bandpass Signals," Proceedings of the IEEE,volume 67, pages 904-919, June 1979.

6. P. Barton, "Digital Beamforming for Radar," IEE Proceedings, volume127, part F, number 4, August 1980.

7. P. D. Carl, G. S. Kino, C. S. Desilets and P. M. Grant, "A DigitalSynthetic Focus Acoustic Imaging System," Acoustic Imaging, volume 8,pp. 39-53, 1978.

8. B. D. Steinberg, "Digital Beamforming in Ultrasound," IEEETransactions on Ultrasonics, Ferro-Electrics, and Frequency Control,volume 39, pp. 716-721, November 1992.

9. Hans Steyskal, "Digital Beamforming Antennas," Microwave Journal,volume 30, No. 1, pp. 107-124, January 1987.

10. R. E. Crochiere and L. R. Rabiner, "Multirate Digital SignalProcessing," Chapter 2, Prentice Hall, 1983.

B. Analog and Hybrid (Analog-Digital) Beamformer Systems

Relevant analog and hybrid (analog-digital) phased array beamformersystem art can be found in the following patents which are incorporatedherein by reference.

    ______________________________________                                        U.S. Pat. No.:                                                                            Title:          Inventor(s):                                      ______________________________________                                        4,140,022   MULTIPLE        Samuel H. Maslak                                              TRANSDUCER                                                                    ACOUSTIC IMAGING                                                              APPARATUS                                                         4,550,607   PHASED ARRAY    Samuel H. Maslak                                              ACOUSTIC IMAGING                                                                              J. Nelson Wright                                              SYSTEM                                                            4,699,009   DYNAMICALLY     Samuel H. Maslak                                              FOCUSED LINEAR  Hugh G. Larsen                                                PHASED ARRAY                                                                  ACOUSTIC IMAGING                                                              SYSTEM                                                            5,014,710   STEERED LINEAR  Samuel H. Maslak                                  and         COLOR DOPPLER   Donald J. Burch                                   5,165,413   IMAGING         J. Nelson Wright                                                              Hugh G. Larsen                                                                Donald R. Langdon                                                             Joel S. Chaffin                                                               Grant Flash, III                                  ______________________________________                                    

C. Digital Beamformer Systems

The concept of a digital receive beamformer system has been proposed inthe art with respect to ultrasound systems. By way of example, thefollowing U.S. patents, all of which are incorporated herein byreference, discuss various aspects of such systems. The patents include:

    ______________________________________                                        U.S. Pat. No.:                                                                          Title:           Inventor(s):                                       ______________________________________                                        4,809,184 METHOD AND       Matthew O'Donnell                                            APPARATUS FOR    Mark Magrane                                                 FULLY DIGITAL BEAM                                                            FORMATION IN A                                                                PHASED ARRAY                                                                  COHERENT IMAGING                                                              SYSTEM                                                              4,839,652 METHOD AND       Matthew O'Donnell                                            APPARATUS FOR HIGH                                                                             WilIiam E. Engeler                                           SPEED DIGITAL    Thomas L. Vogelsong                                          PHASED ARRAY     Steven G. Karr                                               COHERENT IMAGING Sharbel E. Noujaim                                           SYSTEM                                                              4,886,069 METHOD OF, AND   Matthew O'Donnell                                            APPARATUS FOR,                                                                OBTAINING A                                                                   PLURALITY OF                                                                  DIFFERENT RETURN                                                              ENERGY IMAGING                                                                BEAMS RESPONSIVE                                                              TO A SINGLE                                                                   EXCITATION EVENT                                                    4,893,284 CALIBRATION OF   Mark G. Magrane                                              PHASED ARRAY                                                                  ULTRASOUND PROBE                                                    4,896,287 CORDIC COMPLEX   Matthew O'Donnell                                            MULTIPLIER       William E. Engeler                                 4,975,885 DIGITAL INPUT STAGE                                                                            Dietrich Hassler                                             FOR AN ULTRASOUND                                                                              Erhard Schmidt                                               APPARATUS        Peter Wegener                                      4,983,970 METHOD AND       Matthew O'Donnell                                            APPARATUS FOR    WilIiarn E. Engeler                                                           John J. Bloomer                                              DIGITAL PHASED   John T. Pedicone                                             ARRAY IMAGING                                                       5,005,419 METHOD AND       Matthew O'DonneIl                                            APPARATUS FOR    Kenneth B. Welles, II                                        COHERENT IMAGING Carl R. Crawford                                             SYSTEM           Norbert J. Plec                                                               Steven G. Karr                                     5,111,695 DYNAMIC PHASE    William E. Engeler                                           FOCUS FOR COHERENT                                                                             Matthew O'DonneIl                                            IMAGING BEAM     John T. Pedicone                                             FORMATION        John J. Bloomer                                    5,142,649 ULTRASONIC IMAGING                                                                             Matthew O'Donnell                                            SYSTEM WITH                                                                   MULTIPLE,                                                                     DYNAMICALLY                                                                   FOCUSED TRANSMIT                                                              BEAMS                                                               5,230,340 ULTRASOUND       Theador L. Rhyne                                             IMAGING SYSTEM                                                                WITH IMPROVED                                                                 DYNAMIC FOCUSING                                                    5,235,982 DYNAMIC TRANSMIT Matthew O'Donnell                                            FOCUSING OF A                                                                 STEERED ULTRASONIC                                                            BEAM                                                                5,249,578 ULTRASOUND       Sidney M. Karp                                               IMAGING SYSTEM   Raymond A. Beaudin                                           USING FINITE IMPULSE                                                          RESPONSE DIGITAL                                                              CLUTTER FILTER WITH                                                           FORWARD AND                                                                   REVERSE                                                                       COEFFICIENTS                                                        ______________________________________                                    

The basic feature of a digital receive beamformer system as disclosedabove can include: (1) amplification of the ultrasound signal receivedat each element of an array such as, for example, a linear array; (2)direct per channel analog-to-digital conversion of the ultrasound signalwith an analog-to -digital sampling rate at least twice the highestfrequency in the signal; (3) a digital memory to provide delays forfocusing; and (4) digital summation of the focused signals from all thechannels. Other processing features of a receive beamformer system caninclude phase rotation of a receive signal on a channel-by-channel basisto provide fine focusing, amplitude scaling (apodization) to control thebeam sidelobes, and digital filtering to control the bandwidth of thesignal.

This art points out the ever present desire to achieve, in an efficientmanner, a reconstructed image of high quality.

III. SUMMARY OF THE INVENTION

The present invention relates to a method and apparatus for formingsingle or multiple, phase-aligned coherent, steered and dynamicallyfocused receive beams for an ultrasonic imaging system.

A. System Architecture: Independent Parallel Programmable Multi-ChannelDigital Signal Processing Receivers

The present method and apparatus of the invention provides for asubstantially digital signal processing architecture of independentreceivers, preferably assigned one to each available signal from atransducer, which are fully programmable for adjustment of signalparameters and beamformation parameters at rates consistent with dynamicfocusing and with updating at every scan line. Each receiver hasmultiple processing channels that can support formation of multiplesimultaneous beams (scan lines). The independence, programmability, andprocessor channelization support a versatility not available in priorart. The architecture achieves independent receivers (1) by creating aseparate central control apparatus (subject of a co-pending patentapplication) that determines all signal and beamformation parametersindependent of all receivers, and (2) by programming the parameters intoeach receiver at rates needed to sustain dynamic focusing and/orscan-line-to-scan-line adjustments. The digital receive beamformerarchitecture can therefore support conventional beamformation, and canalso support enhanced receive beamformer capability, such as adaptivebeamformation. Signal and beamformation parameters that can beprogrammed on a scan-line interval basis include: delay sample values,apodization sample values, demodulation frequency, signal-shaping filtervalues, gain, sample rate, gain and phase calibration adjustments, andnumber of simultaneous receive beams. The advantage of a systemarchitecture with independent receivers having programmable features isthe ability to support new receive beamformation techniques, which canbe accomplished by reprogramming the types of parameters sent to thereceivers.

B. System Architecture--Maximum Computational Capacity Utilization

The present method and apparatus of the invention provides for anarchitecture having a unique arrangement and implementation of digitalsignal processing and beamforming building blocks which provide enhancedbeam reconstruction. The arrangement of the building blocks as a wholeprovides for a digital receive beamformer system with full and maximumsignal processing computational capacity utilization. This innovativearchitecture allows for processing mode trade-offs among (1) receivesignal nominal center frequency F₀, (2) receive signal spatial rangeresolution γ_(B) (inversely related to receive signal bandwidth) and (3)the number of simultaneously received beams N_(B), for a broad range ofimaging frequencies such that the computational building blocks of thearchitecture are efficiently used to their maximum. Thus for example,when operating at a selected center frequency, the number ofsimultaneously received beams can be traded against the rangeresolution. The digital receive beamformer system can, therefore,operate at a full computational capacity and without having idle processtime or hardware. Advantages of such flexibility include the fact that alarger number of beams affords a higher frame rate which is desirable,for example, when imaging moving objects.

Another aspect of the method and apparatus of the present invention isthat the high rate computations are performed by more blocks of thearchitecture before any distinction is made, in a dynamic delay memory,among the beams of a multiple beam system. This reduces the number ofcomputations required, increases system speed and versatility inaccordance with the utilization of full computational capacity, andminimizes dynamic delay memory size.

C. System Architecture--Versatility

It is to be understood that the various aspects of the method andapparatus of the invention can afford significant advantages when usedby themselves without dependency from other aspects of the invention, asmore fully discussed in the detailed description and outlined in theclaims. By way of example only, the variable time delay memory can beused in other arrangements than specified above. The variable time delaymemory can be used in a system which can include a complex multiplierbut where at least one of the first and second decimators would not bepresent. Also by way of example, the local control processor forproducing delay, apodization, phase and frequency, and calibrationvalues could be used separately or together with some or all of thecontrol values and parameters supplied for the central beamformercontrol as discussed in co-pending patent application entitled: METHODAND APPARATUS FOR FOCUS CONTROL OF TRANSMIT AND RECEIVE BEAMFORMERSYSTEMS.

Along this same concept of independent advantages, it is noted that thetime delay memory can be uniquely used to separate data from amultiple-beam data stream in receive path arrangements other than thatspecified herein.

Accordingly, the present invention improves upon the prior art bydefining a computationally efficient beamformer which allows a trade-offamong (1) receive signal nominal center frequency F₀, (2) normalizedper-beam relative spatial range resolution γ_(B) /λ_(O) (inverselyrelated to receive signal bandwidth), and (3) number of beams N_(B).

Additionally, the digital receive beamformer comprises a means forprocessing a signal which is representative of one or more beams, withthis processing including means for adjusting the spatial rangeresolution (receive signal bandwidth) depending on the number of beamsrepresented by the signal. Further, the adjustment of the receive signalspatial range resolution is also related to the receive signal nominalcenter frequency for each beam.

The system also provides for efficient decimation operations. Decimationin general provides for a signal with reduced data which reducescomputational needs downstream.

Additionally, the present invention provides for dynamically varying thetime delay and apodization value on a sample-by-sample basis for everysample in the scan format. Thus, instead of applying a single time delayand/or apodization value, dynamically varying time delay and/orapodization values provide for a totally reconstructed new data stream.This arrangement has an additional advantage that multiple beams can bereconstructed out of the delay memory in one digital signal path.

D. Arrangements of Per Multi-Channel Processor Digital Signal ProcessingComponents

In another aspect of the invention the post ADC digital receive signalprocessing architecture provides for (1) a first programmable decimator,(2) a dynamic or variable time delay memory, and (3) a secondprogrammable decimator. Such an arrangement affords the above advantagewith respect to full and maximum signal processing computationalcapacity utilization. Thus, the relationship between (1) receive signalnominal center frequency F₀, (2) receive signal spatial range resolutionγ_(B) (inversely related to receive signal bandwidth), and (3) thenumber of simultaneously received beams N_(B) can be implemented by theselection of decimation rates with respect to the decimators and inparticular the second decimator, and with respect to the application oftime delay values to the memory in order to distinguish among receivebeams.

In still another aspect of the method and apparatus of this invention,the system can include a time delay memory followed by a decimator inorder to afford the above computational efficiencies and trade-offs.

In yet another aspect of the invention signal demodulation to or nearbaseband (defined as at or near zero Hertz) can occur in a variety ofdifferent locations and be within the spirit and scope of the inventionwhich affords maximum and flexible computational capacity utilization.By way of example only, demodulation could occur prior to the firstdecimator, after the second decimator, or elsewhere in the signal pathbetween the first decimator and the second decimator.

In another embodiment, the demodulation can occur after the stage whereall of the individual channels of the beamformer processors are summedinto a representation of a receive scan line and prior to detection andvideo image processing. In the below present preferred embodiment,demodulation occurs both as part of the operation of the seconddecimator, and as part of the operation of the complex multiplier.Alternatively, demodulation could occur only as part of the operation ofthe second decimator or only as part of the operation of the complexmultiplier.

In another aspect of the method and apparatus of the invention, thesystem can include a time delay memory, followed by a decimator and acomplex multiplier.

Another novel aspect of the method and apparatus of the invention is thetime delay memory. The time delay memory can selectively and dynamicallychoose from a string of data samples stored in the memory in order totrack receive signals with depth along each receive beam by dynamicallyadjusting the time delay input to the time delay memory. Data samplesfrom each receive analog-to-digital converter can preferably be storedin the memory in the order sampled by the analog-to-digital converter.Selection of those data samples received and stored, first-in-time, inthe memory cause a greater signal delay than the selection of those datasamples received and stored, later-in-time, in the memory. Time delayprofiles, which profiles can as required for image reconstruction changewith range, can be used to selectively and dynamically choose thedesired signal time delay, and thus which signal data is addressed andread out of the time delay memory. Further, multiple time delay profilescan be used to select time delays during multiple beam operation. Thus,through the use of multiple delay profiles, the receive signal datastored in the time delay memory can be separated into multiplerepresentations required for the formation of individual receive beams.

In another aspect of the method and apparatus of the invention, thereceive signal processing architecture provides for (1) a firstprogrammable decimator which provides filtering and decimation, (2) avariable time delay memory, (3) a second programmable decimator whichprovides filtering, decimation and demodulation to or near baseband(i.e. to or near zero Hertz), and (4) a complex multiplier structurewhich can selectively provide for signal phasing and apodization, andwhich can also provide residual demodulation of the signal to or nearbaseband. (Under certain operating conditions, the filter of the firstprogrammable decimator, and the filter of the second programmabledecimator can be selectively operated in a bypass mode.)

In yet another aspect of the method and apparatus of the invention thefirst and second decimator structures are programmable to selectivelyutilize a number of different filter characteristics including filtercoefficients and decimation factors to achieve the desired results.Programmability means that the filter coefficients and decimationfactors may be permanently stored in the filters and decimators, withthe system selecting among such stored values, or these values can bedownloaded from, for example, a central control. Thus, as used herein, adevice or function which is programmable includes those which can beprogrammed either (1) by providing as required a set of specific values(downloaded from, for example, a central control) for use by the deviceor function, or (2) by selecting such values from a pre-determined setof available values which are pre-stored by the device or function.Similarly, the acts of programming carry the same meanings. Suchstructures can include one or more digital filters provided in variousarrangements with a decimator. The second programmable decimator canalso simultaneously provide for demodulation near or to baseband throughthe use of real-or complex-valued filter coefficients.

In still another aspect of the method and apparatus of the invention,decimator two can as suggested above determine the desired receivespatial range resolution (bandwidth mode) for the signal in acomputationally efficient manner.

In yet another aspect of the method and apparatus of the invention, thesecond decimator provides demodulated complex I (In-phase) and Q(Quadrature) data.

Another novel aspect of the method and apparatus of the inventionincludes the complex multiplier which applies a focusing phase shift(derived from focusing delay), apodization, and in a preferredembodiment, demodulation. It is to be understood that the phase shift,apodization, and demodulation can be applied in one combined structure,or in separate structures located adjacent to each other or positionedadvantageously in other locations of the signal path. This phase shiftis in addition to the time delay which is applied to the variable timedelay memory. The focusing phase shift and the apodization values aredynamically variable across the individual elements of the transducerarray and also along the range direction of the receive scan. Thus, thesignal received can be dynamically focused on a sample-by-sample basisfor all data samples in the scan format to form the image.

E. Secondary Control of Multi-Channel Processor Digital SignalProcessing for Receive Beamformation

In another aspect of the invention, the method and apparatus includes alocal or secondary control per receive processor which cooperates with acentral or primary control. The local or secondary control dynamicallyprovides time delay, phase rotation, apodization and calibration valuesto each sample of each receive beam. The local or secondary controlincludes a memory address and delay processor, a phase and frequencyprocessor, an apodization processor, and a calibration processor.

In another aspect of the invention, the memory address and delayprocessor calculates time delay values to be applied to the delaymemory, on a per beam and per sample basis.

In a further aspect, the phase and frequency processor calculates phasevalues which can be a summation of a demodulation phase value and a finefocusing time delay in the form of a phase rotation or shift value atthe nominal center frequency, on a per-beam and per-sample basis. Thephase and frequency processor can track receive signal frequencydownshifting with tissue depth due to attenuation.

In a further aspect, the apodization processor calculates apodizationvalues on a per beam and per sample basis.

In another aspect, the calibration processor calculates calibrationvalues on a per analog receive path sample basis.

Additional advantages, objects, and novel features can be obtained froma review of the specification and the figures.

IV. BRIEF DESCRIPTION OF THE FIGURES

FIGS. 1a and 1b conceptually depict the transmission and reception ofultrasound beams to and from body tissue.

FIG. 2a depicts a high level block diagram schematic of a novelultrasound beamformer system of an ultrasound medical imaging systemincluding an embodiment of a digital receive beamformer system of theinvention.

FIGS. 2b and 2c taken together depict a detailed block diagram of theultrasound beamformer system of FIG. 2a.

FIG. 3 depicts a detailed block diagram of an embodiment of a digitalmulti-channel receive processor and baseband multi-beam processor of theinvention of FIG. 2.

FIG. 4 depicts a schematical representation of the variable time delaymemory of the digital multi-channel receive processor of FIG. 3 of theinvention, along with an embodiment of the memory address and delayprocessor.

FIGS. 5a and 5b depict graphs of typical time delay profiles which canbe applied to the variable time delay memory of FIG. 4.

FIG. 5c depicts a series of evolving delay profiles which haveincreasing aperture widths with increased range along a receive scanline centered on and normal to the transducer array.

FIGS. 6a, 6b and 6c depict graphically the storage and selection ofappropriate time delayed data from the variable time delay memory of thedigital multi-channel receive processor of FIG. 4.

FIG. 7 depicts schematically the selection of data stored in thevariable time delay memory of FIG. 4 for purposes of outputting delaydata representative of that used to form multiple beams.

FIG. 8 depicts a schematic of an embodiment of the complex multiplier,the phase and frequency processor, and the apodization processor of thelocal processor control of the invention.

FIG. 9 is a block diagram schematic of an embodiment of a phase alignerof the invention which provides for phase alignment among receive scanlines in conjunction with a decimator, and a phase aligner (gain, phaseand frequency) control processor.

FIGS. 10a, 10b and 10c depict graphs of typical signal frequencydownshifting profiles that can be applied for signal demodulation andfine phase adjustment in the complex multiplier and for signalremodulation in the phase aligner. FIGS. 10d, 10e and 10f depict graphsof signal frequency downshifting profiles appropriate for signaldemodulation.

FIG. 11 depicts a series of differently evolving apodization profileswhich have increasing aperture widths with increased range along areceive scan line centered on and normal to the transducer array.

V. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention represents a component of a medical ultrasoundimaging system for which additional patent applications, listed above,have been simultaneously filed in the United States Patent and TrademarkOffice.

A. Overview of the Preferred Beamformer System Architecture

1. Ultrasound Signal Description

With respect to the present invention, ultrasound imaging isaccomplished by firing (transmitting) into body tissue or other objectsto be imaged a scan sequence of focused ultrasonic beams centered alongstraight lines in space called transmit scan lines (FIG. 1a). Thetransmit scan lines are generated by a transmit beamformer and anultrasound transducer array. The transmit scan lines are spaced toproduce a planar linear, planar sector or other display of the tissuevia a pre-defined firing or scanning pattern. Focused to some defineddepth in the tissue, the ultrasonic transmit continuous-wave (CW) orpulse-wave (PW) signal, propagating at an assumed constant propagationvelocity of nominally c=1540 m/sec through the tissue, interacts withthe tissue and reflects a small portion of the signal back to theultrasound transducer array that initiated the ultrasound signal. Theround trip delay time is shortest for those targets closest to theultrasound transducer array, and longest for those targets farthest fromthe transducer array. With the application of appropriate time delays,the receive beamformer (FIG. 1b) can dynamically focus receive beamsalong straight lines in space called receive scan lines commencing, forexample, with the shallowest range (depth) of interest and evolvingtoward the deepest range of interest.

FIGS. 1a and 1b depict representations of transmit and receive scanlines (solid) and straight-line signal propagation paths from individualelements (dashed), respectively. In FIG. 1a, the transmit beamformer isgenerally identified by T-50 with the transducer array T-52 containing amultiplicity of individual transducer elements T-54 organized as alinear phased array in this particular embodiment. As is known in theart, there are a great variety of transducer array configurationsavailable for use with ultrasound transmit and receive beamformersystems. As can be seen in FIG. 1a, the transmit beamformer T-50 sendsappropriately time-delayed electrical signals to the individualtransducer elements T-54. These transducer elements T-54 then in turnconvert electrical signals into acoustic waves that propagate into thebody tissue T-56. By applying different time delays to the excitationsignals sent to the individual transducer elements T-54, transmit scanlines T-60 and T-62, having respective foci r₁ and r₂, can beestablished. It is to be understood that each of these transmit scanlines is representative of a center line of a different transmit beamwhich is steered and focused into the body to be imaged.

The transmit beamformer T-50 can generate simultaneous multiple beamsalong different scan lines, or different focal depths along the samescan line (compound focus). Further, the multiple transmit beams caneach scan the entire image format or be transmitted such that each ofthe multiple beams only scans a specified section of the image format.

FIG. 1b depicts a digital receive beamformer R-58 which is alsoconnected to the transducer array T-52. Also depicted in FIG. 1b arereceive scan lines R-64, R-66 corresponding to a dynamically focusedfirst receive beam and a dynamically focused second receive beam,respectively. The beams are sampled in range at a plurality of focaldepths (r₁, r₂, r₃) along each scan line. In the digital receive signalpath of the present invention, transducer array signals can beselectively separated into data representative of multiple individualbeams.

Each scan line of a transmit or receive scan pattern can beparameterized by the origin on the transducer array, the scan lineorientation (angle 0) and the focus depth or range (r). The ultrasoundimaging system of the present invention stores a pre-computed sparsedata set of focusing time delay and aperture apodization values indexedby these parameters (based on geometric considerations as is known inthe art) and expands the values by real-time computational means tocontrol the transmit and receive beamformation systems that produce thedesired scan lines.

2. Beamformer System

FIGS. 2a, 2b, 2c depict an overall block diagram of a medical ultrasoundimaging system R-20. Ultrasound system R-20 includes a beamformer systemR-22, one or more transducers T-112, a display processing system R-26with a display R-28 and an ultrasound imaging system control R-40.

In FIGS. 2a, 2b, or 2c, the beamformer system R-22 includes inventiveand novel (1) digital transmit beamformer system T-102, (2) digitalreceive beamformer system R-100, (3) beamformer central control systemC-104, (4) adaptive focusing control system G-100, (5) Doppler receivebeamformer system A-400, (6) baseband multi-beam processor R-125, and(7) coherent sample synthesizer S-100. These systems are depicted ashigh level, functional block diagrams. The blocks are abstracted fromthe actual implementation of a preferred embodiment in order to betterillustrate the signal processing functions performed.

As indicated in FIG. 2a, beamformer system R-22 provides two sources ofdigital beam data to the display processing system R-26: (1) Dopplerreceive beamformer single-beam complex in-phase/quadrature datarepresenting coherent temporal sampling of the beam (CW case) orcoherent temporal sampling at one range location along the beam (PWcase), and (2) digital receive beamformer multi-beam complexin-phase/quadrature data representing coherent sampling in range alongeach receive scan line. Beamformer system R-22 can be operated toprovide a sequence of scan lines and associated samples as above toprovide data for a variety of display modes. By way of example, possibledisplay modes and their associated processors include (1) brightnessimage and motion processor R-30 for B-mode (gray-scale imaging) andM-mode (motion display), (2) color Doppler image processor R-32 for flowimaging, and (3) spectral Doppler processor R-34 for wide dynamicnonimaging Doppler velocity vs. time displays. Additional display modescan be created from the two complex data sources of R-22, as will beobvious to those skilled in the art.

Ultrasound system R-20 also includes a transmit demultiplexer T-106 forrouting the output waveforms from the transmitters T-103 to thetransducer elements T-114, a receive multiplexer R-108 for routing theinput waveforms from the transducer elements T-114 to the receiversR-101, one or more transducer connectors T-110 and transducer arraysT-112. Many types of transducer arrays can be used with the presentsystem.

Ultrasound system R-20 also includes an ultrasound imaging systemcontrol R-40, archival memory R-38 for storing scan parameters and scandata, and operator interface R-36.

As used herein, the term ultrasonic refers to frequencies above therange of human hearing. However, the transducer arrays T-112 areoptimized for frequencies typically within the range of 2-10 MHz.

The transducer array T-112 is interchangeable with a variety ofdifferent kinds of transducer arrays, including but not limited tolinear, curved, curvi-linear and annular transducer arrays. A variety oftransducer array shapes and frequencies are desirable in order tosatisfy the requirements of a variety of different clinical settings.However, the transducer arrays T-112 are typically optimized forfrequencies within the above specified range of 2-10 MHz. The medicalultrasound system R-20 performs the three major functions of driving theultrasonic transducer array of elements T-114 to transmit focusedultrasound energy, receiving and focusing back-scattered ultrasoundenergy impinging on the transducer array T-114, and controlling thetransmit and receive functions to scan a field of view in scan formatsincluding (but not limited to) linear, sector or Vector® format.

In FIGS. 2a, 2b, 2c, the control signals are communicated over the lightlead lines while the signal paths are depicted with heavy lead lines.

3. Digital Transmit Beamformer System

The digital transmit beamformer T-102 (FIG. 2c) is the subject of theabove cited co-pending application entitled: METHOD AND APPARATUS FORTRANSMIT BEAMFORMER SYSTEM. It is to be understood that in a preferredembodiment, the digital transmit beamformer T-102 is comprised of aplurality of digital multi-channel transmitters T-103, one digitalmulti-channel transmitters for one or more of the individual transducerelements T-114. The transmitters are multi-channel in that eachtransmitter can process, in a preferred embodiment, up to fourindependent beams. Thus, for example, 128 multi-channel transmittershave 512 channels. In other preferred embodiments, more than fourindependent beams can be achieved. Processing more than four beams perprocessor is within the scope of the invention.

In a preferred embodiment, each of the digital multi-channeltransmitters T-103 produces as its output in response to an excitationevent the superposition of up to four pulses, each pulse correspondingto a beam. Each pulse has a precisely programmed waveform, whoseamplitude is apodized appropriately relative to the other transmittersand/or channels, and delayed by a precisely defined time delay relativeto a common start-of-transmit (SOT) signal. Transmitters T-103 are alsocapable of producing CW.

Each digital multi-channel transmitter T-103 conceptually comprises amultiple beam transmit filter T-115 which provides an output to acomplex modulator T-117. The output from complex modulator T-117 iscommunicated to a delay/filter block T-119, and therefrom is provided toa digital-to-analog converter (DAC) T-121. The output of the DAC T-121is amplified by an amplifier T-123. The multiple beam transmit filterT-115, the complex modulator T-117 and the delay/filter block T-119comprise a digital multi-channel transmit processor T-104.

The transmit filter T-115 can be programmed to provide any arbitraryreal or complex waveform responsive to a start-of-transmit (SOT) signal.The transmit filter T-115 is implemented with a memory which stores realor complex samples of any desired and arbitrary pulse waveform, and ameans of reading the samples out sequentially in response to thestart-of-transmit (SOT) signal delayed by a component of the focusingdelay. In a preferred embodiment, the memory of T-115 is programmed tostore baseband representations of real or complex pulse envelopes.

Block T-115, although primarily a memory, is referred to herein as atransmit filter, as the output of block T-115 can be thought of as thetime response of a filter to an impulse. The complex modulator T-117upconverts the envelope to the transmit frequency and providesappropriate focusing phase and aperture apodization.

Delay/filter block T-119 conceptually provides any remaining focusingdelay component and a final shaping filter. The digital-to-analogconverter (DAC) T-121 converts the transmit waveform samples to ananalog signal. The transmit amplifier T-123 sets the transmit powerlevel and generates the high-voltage signal which is routed by thetransmit demultiplexer T-106 to a selected transducer element T-114.

Associated with each multi-channel transmit processor T-104 is a localor secondary processor control C-125 which provides control values andparameters, such as apodization and delay values, to the functionalblocks of multi-channel transmit processor T-104. Each local orsecondary channel control C-125 is in turn controlled by the central orprimary control system C-104.

4. Digital Receive Beamformer System

The digital receive beamformer R-100 (FIG. 2b) is the subject of thethis application.

The signals from the individual transducer elements T-114 representreturn echoes or return signals which are reflected from the objectbeing imaged. These signals are communicated through the transducerconnectors T-110 to the receive multiplexer R-108. Through multiplexerR-108, each transducer element T-114 is connected separately to one ofthe plurality of digital multi-channel receivers R-101 which takentogether with summer R-126 comprise the digital receive beamformer R-100of the invention. The receivers are multi-channel in that each receivercan process, in a preferred embodiment, up to four independent beams.Processing more than four beams per processor is within the scope of theinvention.

Each digital multi-channel receiver R-101 can, in a preferredembodiment, comprise the following elements which are represented by thehigh level function block diagram in FIG. 2b. These elements include adynamic low-noise and variable time-gain amplifier R-116, ananalog-to-digital converter (ADC) R-118, and a digital multi-channelreceive processor R-120. The digital multi-channel receive processorR-120 conceptually includes a filter/delay unit R-122 and a complexdemodulator R-124. The filter/delay unit R-122 provides for filteringand coarse focusing time delay. The complex demodulator R-124 providesfor fine focusing delay in the form of a phase rotation and apodization(scaling or weighting), as well as signal demodulation to or nearbaseband. The digital multi-channel receivers R-101 communicate withsummer R-126 where the signal samples associated with each beam fromeach receive processor are summed to form final receive scan linesamples, and the resulting complex samples provided to basebandprocessor R-125. The exact functioning and composition of each of theseblocks will be more fully described hereinbelow with respect to theremaining figures.

A local or secondary control C-210 is associated with each digitalmulti-channel receiver R-101. Local processor control C-210 iscontrolled by central or primary control C-104 and provides timing,control and parameter values to each said receiver R-101. The parametervalues include focusing time delay profiles and apodization profiles.

5. Doppler Receive Beamformer System

The Doppler receive beamformer system A-400 for wide dynamic range,nonimaging Doppler acquisition includes analog receivers A-402, each ofwhich receives echo signals from a respective one or more transducersT-114. Each of the Doppler receivers A-402 includes a demodulator/rangegate A-404 which demodulates the received signal and gates it (PW modeonly) to select the echo from a narrow range. The analog outputs of theDoppler receivers A-402 are communicated to a Doppler preprocessorA-406. In preprocessor A-406, the analog signals are summed by summerA-408 and then integrated, filtered, and sampled by analog processorA-410. Preprocessor A-406 then digitizes the sampled analog signal in ananalog-to-digital converter (ADC) A-412. The digitized signal iscommunicated to the display processing system R-26. The Doppler receivebeamformer system is the subject of the above identified co-pendingpatent application entitled: METHOD AND APPARATUS FOR DOPPLER RECEIVEBEAMFORMER SYSTEM which has been incorporated herein by reference.

Associated with all Doppler receivers A-402 is a single local orsecondary Doppler beamformer control C-127. Doppler beamformer controlC-127 is controlled by central or primary control system C-104 andprovides control and focusing parameter values to the Doppler receivebeamformer system A-400.

As pointed out in the above patent application describing the Dopplerreceive beamformer system A-400, the present beamformer system R-22advantageously combines an imaging digital receive beamformation systemR-100 and a nonimaging Doppler receive beamformation system A-400 in amanner which uses the same digital transmit beamformation system T-102and the same transducer array and allows the digital receivebeamformation system R-100 to be optimized for imaging modes such asB-mode and color Doppler imaging, and therefore has high spatialresolution, while the accompanying Doppler receive beamformation systemhas a wide dynamic range and is optimized for use in acquiring signalsfor nonimaging Doppler processing.

6. Beamformer Central Control System

The beamformer central control system C-104 of the present inventioncontrols the operation of the digital transmit beamformer system T-102,the digital receive beamformer system R-100, the Doppler receivebeamformer system A-400, the adaptive focusing control system G-100, andthe baseband processor R-127. The beamformer control is more fullydiscussed in the above referenced and incorporated patent applicationentitled: METHOD AND APPARATUS FOR FOCUS CONTROL OF TRANSMIT AND RECEIVEBEAMFORMER SYSTEMS.

The main control functions of the central control system C-104 aredepicted in FIG. 2c. The control functions are implemented with fourcomponents. The acquisition control C-130 communicates with the rest ofthe system including the ultrasound system control R-40 and provideshigh level control and downloading of scanning parameters. The focusingcontrol C-132 computes in real time the dynamic delay and apodizationdigital values required for transmit and receive beamformation, whichincludes pre-computed and expanded ideal values plus any estimatedcorrection values provided by adaptive focusing control system G-100.The front end control C-134 sets the switches for the demultiplexerT-106 and the multiplexer R-108, interfaces with the transducerconnectors T-110, and sets the gain and bias levels of all transmitteramplifiers T-123 and all receive amplifiers R-116. The timing controlC-136 provides all the digital clocks required by the digital circuits.This includes the sampling clocks for all the transmitter DACs T-121 andreceiver ADCs R-118.

In a preferred embodiment central control C-104 expands sparse tables offocusing time delay and aperture apodization values based onpre-computed and stored data, through such techniques as interpolationand extrapolation. The expanded delay and apodization values arecommunicated as a profile of values across the transducer aperture tothe local processor controls, where the delay and apodization dataexpansion in range is completed to per-transducer-element, per-sample,per-beam values.

7. Adaptive Focusing Control System

Adaptive focusing control system G-100 provides for real time concurrentadaptive focusing. Adaptive focusing control system G-100 is comprisedof an adaptive focus processor G-505 which provides focus correctiondelay values to the focus control C-132 of the central control C-104.Adaptive focus processor G-505 operates on output produced by aberrationvalue estimators G-502 from data gathered from the subarray summersR-126 of the digital receive beamformer system R-100. Accordingly,aberration correction values, preferably aberration delay and amplitudevalues, are adaptively measured for each receive scan line or for asubset of receive scan lines in range regions corresponding to transmitfocal depths by the adaptive focusing control subsystem G-100 shown inFIG. 2c. Adaptive focusing control system G-100 is more fully describedin the above identified co-pending patent application entitled: METHODAND APPARATUS FOR REAL TIME, CONCURRENT ADAPTIVE FOCUSING IN ANULTRASOUND BEAMFORMER IMAGING SYSTEM.

It is to be understood that in addition to the adaptive focusing controlsystem which adjusts focus delays, that a number of adaptive controlsystems are contemplated. These systems, by way of example, include (1)adaptive contrast enhancement control system for adjusting focus delaysand aperture apodizations, (2) adaptive interference cancellationcontrol for adjusting focus delays and phases, aperture apodizations,and (3) adaptive target enhancement control for adjusting focus delaysand phase, aperture apodizations, imaging transmit and receivefrequencies and baseband waveform shaping.

Another aspect of adaptive focusing which can be included in thepreferred embodiment of the adaptive focusing control system G-100 is ageometric aberration transform device G-508/509 which can provideaberration correction delay values to the adaptive focus processor G-505for scan lines and scan line depth locations for which measuredaberration values were not collected by aberration value estimatorsG-502. More specifically, measured aberration correction values arewritten to a delay table in G-508/509. G-508/509 retrieves values fromthe delay table according to look-up rules of the geometric aberrationtransform to form focusing delay correction profiles across the aperturevalid for depths, scan geometries, and acquisition modes other than thedepth, scan geometry, and mode for which aberration correction valueswere measured. The geometric aberration transform device G-508/509 isthe subject of the above identified co-pending U.S. patent applicationentitled: METHOD AND APPARATUS FOR A GEOMETRIC ABERRATION TRANSFORM INAN ADAPTIVE FOCUSING ULTRASOUND BEAMFORMER SYSTEM.

8. Baseband Processor System The baseband processor R-125 provides forfiltering, and receive-scan-line-to-receive-scan-line (beam-to-beam)amplitude and phase adjustments as discussed herein and in theabove-referenced and incorporated patent applications entitled: METHODAND APPARATUS FOR A BASEBAND PROCESSOR FOR A RECEIVE BEAMFORMER SYSTEMand METHOD AND APPARATUS FOR ADJUSTABLE FREQUENCY SCANNING IN ULTRASOUNDIMAGING, and the above-referenced patent application entitled METHOD ANDAPPARATUS FOR COHERENT IMAGE FORMATION.

The baseband processor R-125 additionally includes a baseband filter, acomplex multiplier, and a baseband processor control which controls theoperation of the baseband filter and complex multiplier. The basebandprocessor control is controlled by central control C-104.

9. Coherent Sample Synthesizer System

The coherent sample synthesizer system S-100 (FIG. 2a) is the subject ofthe above-identified application entitled: METHOD AND APPARATUS FORCOHERENT IMAGE FORMATION.

This system exploits the multi-beam transmit and multi-beam receivecapability of the invention to acquire and store coherent(pre-detection) samples of receive beam data along actual scan lines andto perform interpolation of the stored coherent samples to synthesizenew coherent samples at new range locations along existing scan lines oralong synthetically-created scan lines. Both acquired and synthesizedsamples are passed to the display processing system R-26.

10. Transmit and Receive Multiplexers

The connectivity between the transducer array elements T-114 and theprocessors T-103, R-101, A-402 of the digital transmit, digital receive,and Doppler receive beamformer systems is established through a transmitdemultiplexer T-106 and a separate receive multiplexer R-108, as shownin FIG. 2a. The multiple-transducer multiplexer configuration shown inFIG. 2a permits selection of transmit and receive apertures lyingentirely within a single transducer array or straddling across twotransducer arrays. The two multiplexers are independently controlled bythe beamformer central control system C-104 and may be programmed tosupport a number of acquisition modes, including sliding aperture andsynthetic aperture modes. The multiplexers and their connectivity arethe subject of the above-cited copending application entitled: METHODAND APPARATUS FOR BEAMFORMER SYSTEM WITH VARIABLE APERTURE.

B. Digital Receive Beamformer System Preferred Embodiment

1. Analog Front End

a. Low Noise, Variable Time-Gain Amplifier

As is known in the art, a time-varying gain is applied to the receivesignal to compensate for attenuation with depth. In this embodiment, thegain is applied by an analog low noise, time-gain amplifier R-116 (FIG.2b). There is one low noise, time-gain amplifier R-116 for each digitalmulti-channel receiver R-101. A common gain function is applied to allamplifiers R-116, although independent gains could be applied to eachamplifier R-116. The gain varies with the range (or time, as range andtime are related to each other in accordance with the speed of sound inthe medium being imaged) from the object being imaged to the transducerelements.

b. Analog-To-Digital Converter (ADC)

The analog-to-digital converter (ADC) R-118 (FIG. 2b) in the preferredembodiment oversamples the signal by at least four times (preferablyfour, eight, sixteen or thirty-two times) the receive signal nominalcenter frequency F_(o). It is to be understood that the oversample ratecan be lower or greater than four times and remain within the spirit andscope of the invention. Thus, if the system is imaging at 10 MHz, theADC R-116 is sampling at a rate of 40 MHz. Preferably the ADC R-116 isan eight or more bit ADC. However, it is to be understood that as isevident from the patents listed before, many types of ADCs can be usedwith the beamformer and be within the scope of the invention.

2. Multi-Channel Digital Signal Processing (Digital Multi-ChannelReceive Processor R-120)

a. Processing Modes

Before describing the functional blocks in FIG. 3, it will be useful tounderstand the various processing modes in which each receive processorcan operate. Ideally, it would be desirable for each receive processorto be able to process any number of superposed and separately delayedand apodized receive beams up to some maximum, at any receive signalnominal center frequency F₀ up to some maximum, specified by a receivesignal spatial range resolution γ_(B) (inversely related to receivesignal bandwidth) up to some maximum. This would require extensiveprocessing power, however, especially if the maximums are large. Sinceprocessing power is limited in any system, it would seem that thesemaximums must be kept low enough such that the hardware is able to keepup when all three parameters are specified at maximum. The presentembodiment, on the other hand, makes better use of the availableprocessing power by permitting trade-offs among these three parametersand allowing the central control system to choose among processing modesdepending on the clinical setting. It is to be understood that once theuser selects a transducer, a mode and scan format pursuant to theclinical setting, that preferably the method and apparatus automaticallyselects from the preselected and pre-stored processing modes.

Table I sets forth some of the processing modes which can be selected bycentral control system C-104 to be applied to all digital multi-channelreceive processors R-120 of receive beamformer R-100. Differentembodiments can support fewer or greater numbers of modes and fewer orgreater numbers of beams. As used in the Table:

F_(s) : is the system clock frequency. The central control C-104 can setF, at any of a variety of frequencies.

F_(ADC) : is the ADC sampling frequency or the rate at which samples areconverted by the ADC R-118 (FIG. 2b), where typically F_(ADC) =F_(s) orF_(s) /2.

F₀ : is a receive signal nominal center frequency. F₀ is equal to, ornear, the actual signal carrier frequency F_(c) and is thereforeconsidered to be the nominal receive signal frequency. F_(o) isspecified for each digital multi-channel receiver R-101 as a fraction ofF_(s). F₀ is programmable by the central control C-104 for each digitalmulti-channel receiver R-101 based on prestored values.

c: is the speed of sound in the body.

λ₀ : is the acoustic wavelength of F₀ ; λ_(o) =c/F₀.

F_(c) : is the receive signal carrier frequency (an imaging frequency).The digital multi-channel receiver R-101 can be tuned by verniering. F₀to F_(c). F_(c) and F_(o) are related in the invention by a frequencyscaling factor or frequency vernier factor v, such that v·F_(o) =F_(c)as pre-stored in the central control. The range of the carrierfrequencies F_(c) for which the invention can be tuned rangestheoretically between 0×F_(o) to 2×F_(o), but typically is 75% of F_(o)to 125% of F_(o).

R_(o) : is the per-beam complex (I/Q-pair) output sampling rate or perbeam processing rate. The ratio R₀ /F₀ represents the number of complexsamples per period of the receive signal nominal center frequency F₀.

γ_(B) : is the per-beam spatial range resolution. Note that γ_(B)=c/2R_(o) =λ₀ /(2R_(o) /F₀)

Spatial Range Resolution (or bandwidth modes (BW Mode)) selected atDecimator Two: There are 6 spatial range resolutions (or bandwidthmodes) in the preferred embodiment, accounting for spatial rangeresolution between values F₀ /2 to 4F₀. Values outside these values arewithin the spirit and scope of the invention.

Spatial range resolution (Bandwidth Modes)

BW MODE 1: R₀ =4F₀ or γ_(B) =λ₀ /8.

BW MODE 2: R₀ =2F₀ or γ_(B) =λ₀ /4.

BW MODE 3: R₀ =F₀ or γ_(B) =λ₀ /2.

BW MODE 4: R₀ =F₀ /2 or γ_(B) =λ₀.

BW MODE 5: R₀ =2F₀ /3 or 3γ_(B) =λ₀ /4.

BW MODE 6: R₀ =F₀ /3 or 3γ_(B) =λ₀ /2.

N_(B) =maximum number of simultaneously produced beams for the givenprocessing mode. (Note that the beamformer can be operated to producefewer than N_(B) beams if desired; for example, in a mode for whichN_(B) =4, the beamformer can be operated to produce only three beams ifdesired, although this would not make full use of the available hardwareprocessing power.)

N/I=Mode not implemented in preferred embodiment.

                  TABLE 1                                                         ______________________________________                                        RECEIVE PROCESSING MODES                                                      (Output of Decimator Two)                                                     F.sub.0 (MHz)                                                                           N.sub.B = 1 N.sub.B = 2                                                                              N.sub.B = 4                                  ______________________________________                                        F.sub.s /32                                                                             N/I         BW Mode 1  BW Mode 2                                                          R.sub.o = 4F.sub.0                                                                       R.sub.o = 2F.sub.0                                                 γ.sub.B = λ.sub.0 /8                                                        γ.sub.B = γ.sub.0 /4             F.sub.s /16                                                                             BW Mode 1   BW Mode 2  BW Mode 3                                              R.sub.o = 4F.sub.0                                                                        R.sub.o = 2F.sub.0                                                                       R.sub.o = F.sub.0                                      γ.sub.B = λ.sub.0 /8                                                         γ.sub.B = λ.sub.0 /4                                                        γ.sub.B = λ.sub.0 /2            F.sub.s /8                                                                              BW Mode 2   BW Mode 3  BW Mode 4                                              R.sub.o = 2F.sub.0                                                                        R.sub.o = F.sub.0                                                                        R.sub.o = F.sub.0 /2                                   γ.sub.B = λ.sub.0 /4                                                         γ.sub.B = λ.sub.0 /2                                                        γ.sub.B = λ.sub.0               F.sub.s /4                                                                              BW Mode 3   BW Mode 4  N/I                                                    R.sub.o = F.sub.0                                                                         R.sub.o = F.sub.0 /2                                              γ.sub.B = λ.sub.0 /2                                                         γ.sub.B = λ.sub.0                          3F.sub.s /8                                                                             BW Mode 5   BW Mode 6  N/I                                                    R.sub.o = 2F.sub.0 /3                                                                     R.sub.o = F.sub.0 /3                                              γ.sub.B = 3λ.sub.0 /4                                                        γ.sub.B = 3λ.sub.0 /2                      ______________________________________                                    

As can be seen by reading horizontally across Table 1, for each receivesignal nominal center frequency F₀, the hardware permits a larger numberN_(B) of superposed beam waveforms to be traded-off against somedegradation of the per-beam spatial range resolution γ_(B), andvice-versa. A larger N_(B) translates into a higher frame rate (sincethe entire field of view can be scanned with only half or one quarterthe number of firings), while an enhanced spatial range resolution γ_(B)(smaller value of γ_(B)) translates into a sharper image in range. Forexample, therefore, in a display mode which displays a color flowDoppler image superimposed on a grey-scale image, produced byinterleaving B-mode and F-mode pulse firings, the central control C-104could reprogram the receive beamformer R-100 to operate at N_(B) =1 forall B-mode imaging pulses and at N_(B) =2 or even N_(B) =4 for colorflow Doppler imaging pulses, assuming both modes share the same F₀.

Similarly, reading vertically down Table 1, and excluding modes 5 and 6for this example, it can be seen that for a given maximum number ofbeams N_(B), processing modes having a higher carrier frequency(approximately F₀), have a larger relative per-beam spatial rangeresolution γ_(B). A clinician typically selects a transducer operable atthe carrier frequency appropriate for a desired depth penetration. Indoing so, the clinician trades off penetration for overall imageresolution (ability to distinguish two targets). (The latter trade-offis inherent in the physics of ultrasound since greater penetration isachieved by reducing the imaging frequency, which in turn reduces theoverall image resolution.) For a given maximum number of beams N_(B),the desired tissue penetration determines F₀ (Table 1), which in turndetermines a processing mode having the optimum per-beam spatial rangeresolution which the hardware can provide at the selected F₀. That is,as F₀ decreases relative to F_(s) to achieve greater penetration, thesignal processing path in each receive channel R-101 need not process asmany samples per second. This leaves hardware processing capacityavailable, which the system takes advantage of by increasing R₀ /F₀ andhence improving the normalized per-beam relative spatial rangeresolution γ_(B) /λ₀.

Further, by reading diagonally across Table 1 (upward to the right), andagain excluding modes 5 and 6 for this example, it can be seen that thehardware permits a lower F₀ to be traded off for a larger number ofbeams N_(B) at a constant receive spatial resolution γ_(B).

In summary the modes with which the receive channel R-101 can bespecified to operate offer trade-offs among three parameters: N_(B), F₀,and γ_(B). Thus each processing mode defines a parameter set {N_(B), F₀,γ_(B) }. In general, all of the processing modes shown in Table 1satisfy the rule that for a given F_(s), the product of the maximumnumber of beams N_(B) and the channel processing rate F₀, divided by thenormalized per-beam spatial range resolution γ_(B) /λ₀, is constant.Further, the preferred embodiment also supports additional processingmodes not shown in Table 1, and which do not fully utilize the totalprocessing capability of the system.

b. Decimator One

As can be seen in FIG. 3, the beamformer processor R-120 is comprised ofdecimator one R-150, time delay memory R-152, decimator two R-154 andcomplex multiplier R-156. Decimator one R-150 is programmable (aspreviously defined) and is also referred to as a variable rate decimatorfilter or a multi-rate decimator filter with a variety of programmabledecimation factors and associated programmable filter coefficients.Decimator one R-150, in a preferred embodiment, is functionallycomprised of a first filter (filter one) R-160 which has firstprogrammable filter coefficients hi, a decimator R-162 whichdown-samples at a decimation factor of K_(D1) (Table 2), and a secondfilter (filter two) R-164 which has second programmable filtercoefficients of h2. In a preferred embodiment filter one (h1) is a FIR(finite impulse response), anti-aliasing low/high-pass filter. Filterone (h1) filters out the ADC guantization noise and odd harmonics of thereceive signal nominal center frequency F₀. Preferably, filter two (h2)is a FIR, anti-alias, band-pass filter which filters out the evenharmonics of the receive signal nominal center frequency F₀. The filterprofiles and decimation rate values are programmable depending upon thereceive signal nominal center frequency F₀ and the ADC sampling rate(FADC) Such filters can perform the additional programmable task ofsignal shaping.

In implementation, the functional features of the filter one (h1) R-160and the decimator R-162 are accomplished simultaneously. It is to beunderstood, however, that the filtering and decimating operations canoccur separately and in a less computationally efficient order in otherembodiments and be within the spirit and scope of the invention.

Further it is to be understood that the present invention can beimplemented with filters with a variety of lengths and using fixed orfloating point operations.

A digital signal processing decimator performs both filtering anddownsampling, as described in Sections 2.3.2 and 2.4 of the test byCrochiere and Rabiner, Multirate Digital Signal Processing, PrenticeHall 1983. Decimator filter design is disclosed in Crochiere and Rabinerand in Digital Signal Processing Applications Using the ADSP-2100Family, volume 1, edited by Amy Mar of Analog Devices, DSP division,Prentice Hall 1992, which are hereby incorporated by reference.

In accordance with the same definition of programmable, the programmingof filters and filter coefficients and decimation rates is accomplishedby the central control C-104 which coordinates the operation of thedigital multi-channel transmitter T-103 and the digital multi-channelreceivers R-101. Such filter coefficients and filter values anddecimation factor values can be downloaded to memory R-165 of decimatorone R-150 from the central or primary control C-104. Accordingly,primary control C-104 can program memory R-165 and can select from thevalues programmed into memory R-165 in order to operate decimator oneR-150. Alternatively such values can be permanently pre-stored in amemory such as memory R-165, with the primary control C-104 selectingamong the pre-stored values depending upon the processing mode inaccordance with the above definition of programmable. Further,decimation factors other than those specified in Table 2 can be selectedand allow decimator one to operate within the spirit and scope of theinvention.

According to the Nyquist sampling rule, a real signal must be sampled byat least a factor of two over the highest frequency of the signal inorder to be able to reconstruct the signal successfully. For the signalswhich are received by the digital multi-channel receive processor R-120,there is a significant frequency content above the signal nominal centerfrequency F₀, and at an oversample rate of four times F_(o) (See Table2), these frequencies are adequately sampled. In a preferred embodimentif the data from the ADC R-118 is already at four times F_(o), nodecimation is performed. Thus, one of the normal decimation modes ofdecimator one R-150 is that decimator one R-150 does not decimate atall. With a beam having a signal center frequency F₀ =F₀ of 10 MHz, andwith a sampling frequency F_(s) of 40 MHz, then the output of decimatorone R-150 without decimation would be 40MHz, or four times oversampled.Data from the ADC R-118, which is sampled at greater than four times thereceive signal nominal center frequency F₀, is downsampled to four timesthe receive signal nominal center frequency 4F₀, as is evident fromTable 2. The decimation factors K_(D1) are selected to accomplish thisrate of decimation as a function of the ADC sampling rate F_(ADC).

Accordingly, in this embodiment, the relationship between the decimationfactor K_(D1) for decimator one and the channel processing rate orcenter frequency F₀ and the ADC sampling rate F_(ADC) is K_(D1) =F_(ADC)/4F₀ where F_(ADC) =F_(s) or F_(s) /2.

It is to be understood that oversampling by less than or greater than afactor of 4 (and thus with different integer and/or rational decimationfactors K_(D1)) can be accomplished by this present invention and bewithin the scope of this invention.

Further, for the filter one (h1) R-160 and the filter two (h2) R-162 thefilter coefficients can be selected in order to cause these filters tooperate in a bypass mode (i.e., without filtering) for each of thespecified decimation factors. Such bypass operation may be utilized fordiagnostic purposes. Additionally for maximum wide-band processing,filter one can perform no filtering.

                  TABLE 2                                                         ______________________________________                                        DECIMATION FACTORS FOR DECIMATOR ONE                                                  K.sub.D1 Decimation                                                                              Decimator One                                      F.sub.0 Factor             Output Rate                                        ______________________________________                                        F.sub.s /32                                                                           8                  4F.sub.0                                           F.sub.s /16                                                                           4                  4F.sub.0                                           F.sub.s /8                                                                            2                  4F.sub.0                                           F.sub.s /4                                                                            1                  4F.sub.0                                           3F.sub.s /8                                                                           2                  4F.sub.0 /3                                        ______________________________________                                    

C. Time Delay Memory

As can be seen in FIG. 5a, the time delay profile across the aperture ofa transducer is a function of both the transducer element position andthe range of the object to be imaged from the transducer array.Generally, for the case where the scan line is steered straight ahead,more delay is applied in the center of the aperture (FIG. 5a) than isapplied to the signals at the edges of the transducer array. This is dueto the fact that it takes longer for the receive (return echo)ultrasound signals from the object to be imaged to reach the outertransducer elements than to reach the more central transducer elementsor elements closer to the object to be imaged.

Also as shown in FIG. 5a for the case where the scan line is steerednormal to the transducer array face, the reason that the time delayprofiles are flatter as a function of range (or time to the object to beimaged) is that as the range increases to infinity, the distances fromany particular transducer element to the object to be imaged converge toequal values reducing the need for time delays in order to properly sumthe receive signals.

In a preferred embodiment, different time delay profiles are assigned toreference range boundaries of range zones (FIGS. 5a and 5c and asexplained below). The spacing between the reference range boundaries maybe equal and/or unequal as desired. Further, it is to be understood thatthese time delays represent a coarse time delay applied to the signal asexplained below, with a fine focusing time delay implemented as a phaseshift applied by the complex multiplier R-156 (FIG. 3).

Tracking receive beams that are steered relative to the transduceraperture is a matter of changing the time delay profile with respect tothe number of the transducer element and the range, as can be seen inFIG. 5b. Thus, by changing the time delay profile which is applied toselect time-indexed receive data from memory, the desired beams can besteered and focused.

FIG. 4 depicts a schematic of the programmable, variable time-delay,two-port memory R-152 of the preferred embodiment. Data is read out ofthe memory R-152 based on continuously updated addresses derived fromvariable time delay profiles (such as for example described above),supplied by the central control system C-104 and the local controlprocessor system C-210, in order to provide dynamic focusing.

Shown in FIG. 4 are data-in line R-190 and data-out line R-159 as wellas in-address line C-194 and out-address line R-196. The in-address lineC-194 is updated at a constant rate with a modulo counter C-198. Theout-address R-196 is variable and is comprised of a combination of thein-address less a coarse time delay component of the time delay which issupplied by the central control system C-104 and the local controlsystem C-210. In a preferred embodiment the coarse time delay representsthe most significant bits (MSB) and the fine time delay represents theleast significant bits (LSB) of a time delay word from the local controlsystem C-210. In the preferred embodiment for bandwidth modes 1 to 4 andwith T₀ =1/F₀, the coarse time delay represents integer units of quartercycles (T₀ /4) of the receive signal nominal center frequency F₀ and thefine time delay (phase shift) represents a fractional value of a quartercycle. For Bandwidth Modes 5 and 6 the coarse time delay representsinteger units of three quarter cycles (3T₀ /4) and the fine phase shiftrepresents fractional values of three quarter cycles.

The memory R-152 is organized as a circular buffer which writes over theoldest stored data. The memory does not hold data for the entire scan orreceive line, but just enough data to satisfy the span between theminimum and the maximum time delay that could be applied in order toselect stored signal data. Thus, the necessity of having a much largermemory to store all the data from a scan line is avoided. In a preferredembodiment, the memory for each channel captures the most recent 256data samples along a scan line at a rate of 4F₀. The 256 data samplescorrespond, in a preferred embodiment, to a total delay range of 256x T₀/4=64T₀ for Bandwidth Modes 1 to 4 and a total delay range of 256 ×3T₀/4=192T₀ for Bandwidth Modes 5 and 6.

In FIGS. 6a, 6b and 6c, strings of data stored at times t_(k-1), t_(k),and t_(k+1) are depicted for data on three receive channels for adjacenttransducer elements (N-1, N, N+1). The FIGS. 6a, 6b and 6c thusrepresent a snapshot of the stored signals from three transducerelements frozen in time for the three specified times. Applying theappropriate time delay value along the time axis of the figures selectsthe desired data from the string of stored data. Dynamic focusingresults from real time selection of time delay values in order todetermine the data to be read out of the memory R-152. FIGS. 6a, 6b and6c depict read out of samples of S3, S4 and S5 from the data sequencesstored at time t_(k) from the three channels at the selected timedelays. Thus, the ability exists to dynamically select from the storeddata samples according to the different time delay values in order toprovide for dynamic focusing.

As can be seen in FIG. 7, applying different time delay profiles to thesame data stored in the memory R-152 allows the receive beamformerprocessor R-120 to track and, as depicted, form two receive beams fromthe receive signals at each element.

More particularly, FIG. 7 schematically represents the manner thatmultiple beam data is selected from and read out of the memory R-152.Essentially interleaved time delay values from two or more time delayprofiles at each desired range are applied to the same data stored inthe memory R-152. Each time delay profile causes data corresponding to adifferent beam directed in a different direction to be retrieved fromthe memory and output over the data-out line R-192. Thus, theappropriate selection of time delay profiles causes data to be focusedfor different beams.

More particularly, FIG. 7 depicts a phased array transducer R-112 withtransducer elements N-5 to N+4, R-114. Schematically, sequences of dataR-200 (such as depicted in FIGS. 6a, 6b and 6c) which are stored inmemory R-152 for each transducer element at time "t" are shownassociated with the respective elements. Superimposed over the sequencesof data are first and second time delay profiles R-202, R-204representing profiles for first and second beams (BM₁, BM₂). Byselecting the appropriate time delay values for each transducer elementfrom the time delay profiles for each beam (as provided by the centraland local control system), individual focal points R-206 of first andsecond beams can be formed from the appropriate data from each datasequence.

It is to be understood that the time delay profile can be dynamicallychanged for every instance in time. Thus, any desired beam which iscontained in the data can be tracked and formed out of the data storedin memory R-152.

Further emphasizing the computational flexibility of this digitalreceive beamformer system and referring to Table 1, if it is assumedthat a single beam has a nominal center frequency F₀ of 10 MHz, with asampling rate F_(S) of 40MHz, then only one dynamically focused beamcould be formed with a λ₀ /2 spatial range resolution (BW Mode 3). If,however, the beam had a center frequency of 5MHz, then there issufficient computational bandwidth in the system such that two beams canbe formed with λ₀ /2 spatial range resolution (BW Mode 3). In apreferred embodiment, up to four time-interleaved data streams can becreated from the data stored in memory R-152 by applying four sets ofindependent time delay profiles, one set for each beam. Other prior artsystems are not as flexible and require a separate beamformer for eachadditional beam that is to be formed from data from the same transducerelement. Such prior art systems do not have the ability to applycompletely independent delay, phase and apodization values on asample-by-sample basis for either single or multiple receive beams.

A further key advantage of this architecture is that up to and throughthe storage of receive signal data in the memory R-152, no distinctionor segregation in the data is made between beams. Thus, all of the frontend processing and amplification, the ADC operation and the computationsby the decimator one, all of which are very computational intensive, aswell as the process of storing data in the memory R-152 is donetransparent to the number of beams in the receive signal. Were multiplebeams individually tracked and identified earlier in the signalprocessing chain, then the computations in the decimator one, forexample, would need to be run at a multiple of the number of beams timesthe present sampling rate. Thus, the present system affords asubstantial hardware savings by not distinguishing between beams untilthe data is read out of memory R-152, and by efficient and maximum useof the computational capacity by a trade-off among the number of beamsN_(B) processed, the receive signal nominal center frequency F₀ for eachbeam, and the normalized per-beam relative spatial range resolutionγ_(B) /λ0.

d. Decimator Two

The second decimator, decimator two R-154, is programmable and has afilter and decimation structure (variable rate decimation filter) thatis similar to decimator one R-150, but uses programmable complex filtercoefficients h3 for the third filter R-167. The third filter acts as ananti-aliasing, complex band-pass filter and selects the positive imagefrequencies, and filters out negative image frequencies and out-of-bandnoise. This process of filtering and decimating in R-154 can also, in apreferred embodiment, demodulate the signal to or near baseband andconvert the signal to a complex quadrature signal pair of I (in-phase)and Q (quadrature).

As discussed below, with respect to the preferred embodiment the dataoutput from decimator two represents data from one, two or four beams,with the data representing two or four beams being time interleaved. Asdemonstrated in the Tables 1, 2 and 3, decimator two R-154 is where thereceive sample bandwidth trade-off becomes most evident and the spatialrange resolution is finally determined through the selection of thedecimation factor K_(D2).

Memory R-171 (FIG. 3) is programmable (as the term programmable isdefined above) by central control C-104 with multiple complex filtercoefficients and multiple decimator factors. The filter coefficients anddecimator factors are programmed by the central control C-104 inaccordance with the particular imaging task to be accomplished in thedigital multi-channel receiver.

                  TABLE 3                                                         ______________________________________                                        DECIMATION FACTORS FOR DECIMATOR TWO                                                        K.sub.D2 Decimation                                                                          Decimator Two                                    Decimator Two Modes                                                                         Factor         Output Rate R.sub.o                              ______________________________________                                        BW Mode 1     1              4F.sub.0                                         BW Mode 2     2              2F.sub.0                                         BW Mode 3     4              F.sub.0                                          BW Mode 4     8              F.sub.0 /2                                       BW Mode 5     2              2F.sub.0 /3                                      BW Mode 6     4              F.sub.0 /3                                       ______________________________________                                    

The relationship of the decimation factor of decimator two to thenominal center frequency F₀ defines the output sampling rate R_(o) asset out in Table 3 where K_(D2) =4F₀ /R_(o) for Bandwidth Modes 1 to 4and where K_(D2) =4F₀ /3R_(o) for Bandwidth Modes 5 and 6.

Accordingly, it is evident that as the decimation factor goes down to asmaller value, the sample rate per beam increases with the decimator twoR-154 working at a constant full maximum capacity in all situations.Thus, this preferred embodiment uses decimator two R-154 in order tokeep the computational rate at a maximum constant.

It is to be understood that the bypass modes of decimator two, as fordecimator one, enables the isolation of decimator two for diagnosticpurposes and/or when a signal with a wider bandwidth is desired. By wayof example, for Bandwidth Mode 1, decimator two R-154 can be bypassed.Further, decimator two R-154 can be operated simply as a downsamplerwithout performing a filtering operation.

From the above, it is evident that the beamformer processor R-120decimates the signal to the lowest rate for maximum computationalefficiency consistent with the number of beams utilized and spatialrange resolution requirements.

Thus, it is evident that the above receive signal processingarchitecture provides for (1) a variable time delay memory, and (2) asecond programmable decimator which affords the above advantage withrespect to full and maximum signal processing computational bandwidthutilization. The relationship among (1) receive signal nominal centerfrequency F₀, (2) receive signal spatial range resolution γ_(B), and (3)the number of simultaneously received beams N_(B), can be programmedwith decimation factors with respect to the decimators and in particularthe second decimator, and with respect to the application of time delayvalues to the memory in order to distinguish between beams. Suchadvantages are independent of where signal demodulation occurs in thesignal path.

e. Complex Multiplier

Complex multiplication to handle the complex phase rotation for finetime delay is very computational intensive; however, at this point inthe signal path the signal is decimated down to the lowest sample ratein the signal path, and thus complex multiplication can be handled veryefficiently.

The complex multiplier R-156 accomplishes true complex multiplicationwith a cross-multiplication as explained below.

In the complex multiplier R-156 signal demodulation to or near basebandoccurs in order to account for verniering of F_(o) to F₀. However, asexplained above such demodulation to or near baseband, when for examplethere is no verniering of F_(o), can occur at other locations in thesignal path, such as decimator two, and be within the spirit and scopeof the invention.

In the complex multiplier R-156, a weighting term which is a function ofthe apodization value and the focusing phase shift (corresponding to afine time delay) is multiplied by the signal input from decimator twoR-154. The apodization value and the phase shift value can changedynamically on a sample-by-sample, per receive processor, per beambasis. Thus, these values can dynamically vary across the aperture ofthe transducer as well dynamically vary in time (See FIGS. 5a, 5b, 5cand 11). These values are supplied by the central control system C-104,which is the subject of the above referenced patent application, and thelocal processor control C-210.

In FIG. 3, the preferred embodiment of the complex multiplier R-156 isconceptually shown with a complex I/O signal sample multiplied inmultiplier R-210 by a complex phase value and real apodization valuewhich are combined in a complex multiplier R-260. The complex multiplierR-210 is preferably accomplished by four real multiplication operationsperformed by a time shared Booth multiplier. Alternatively a separatephase multiplier and a separate apodization multiplier can be used inorder to focus the signal. In yet another embodiment, the separate phasemultiplier can be implemented with a Cordic multiplier, and the separateapodization multiplier can be implemented by a Booth multiplier.

The output of the complex multiplier R-156 is represented as follows:

    Y=Acosφ·I-Asinφ·Q+j (Acosφ·Q+Asinφ·I)

where I+jQ is the input channel sample signal to complex multiplierR-156, A is the apodization value and φ is the phase shift value.

It is evident from the above and in particular with respect to thememory R-152 and complex multiplier R-156, that the present inventionimplements true dynamic focusing and dynamic apodization as each datasample per beam per receive processor can be modified dynamically withdelay values, phase values and apodization values as supplied by thecentral control system and local processor control systems. Thus, thepresent invention is capable of using instantaneous delay, phase andapodization values calculated by the central control system for everydata sample.

As indicated above, the complex multiplier as well as the rest of thefunctional blocks of FIG. 3 are preferably implemented in high speeddigital hardware. It is within the spirit and scope of this invention,however, that such functional blocks as, for example, for the complexmultiplier, can be implemented in software with general purposemicroprocessors and in a different computational order and withdifferent algorithms other than specified above. By way of example only,in the complex multiplier the apodization value could be multipliedafter the complex I and Q multiplication occurs. Further, the prior artdescribes other methods of implementing a complex multiplier.

f. Focusing Filter

In another embodiment, the fine focusing delay can also be accomplishedwith a delay interpolator, such as a linear interpolation between thetwo samples closest to the desired delay. A generalization of the delayinterpolator is a focusing filter, as described for filter-and-sumbeamforming in section 6.2.5 of the text by Dudgeon and Mersereau(Multichannel Digital Signal Processing, Prentice Hall, 1985). Such afilter is programmed differently for each digital multichannel receiveprocessor, and each waveform associated with each beam within a receiveprocessor, to account for the desired signal-delay-versus-frequencycharacteristic needed to support receive beamformation. The filter willtherefore generally have a nonlinear phase response. The focusing filtercharacteristics therefore contrast with the signal path filtersassociated with the decimation and demodulation operations whichpreferably have linear-phase responses (therefore yielding no distortionof signals in a filter's pass band) and which are typically set toidentical characteristics in all receive processors. The decimator anddemodulation operation filters are used for waveform shaping, notbeamforming, and the same waveform (with appropriate delay andapodization) is normally created in all receive processors, although theinvention supports selection of different filter characteristics amongreceive processors.

3. Per Channel Local Processor Control System

Secondary or local processor control C-210 (FIG. 3) for the digitalmulti-channel receiver R-101, receives control data from the primary orcentral control C-104. The secondary or local processor control C-210includes a controller and I/O processor C-260, a calibration processorC-262, a memory address and delay processor C-264, a phase and frequencyprocessor C-266, and an apodization processor C-268.

The local processor control C-210 is responsible for providing to thedigital multi-channel receive processor R-120 frequency values (i.e.demodulation frequency, phase correction frequency, and receive signalnominal center frequency F₀, delay values, phase shift values,apodization values and calibration values per digital receive sample andper beam as discussed in detail below). The central control systemC-104, as discussed in the above-referenced patent application, isresponsible for providing to the local processor control C-210 thefollowing: (1) filter coefficient programming (in line with thedefinition of programmable above), decimation factor programming, andcalibration value programming per imaging mode, (2) frequency parametersas specified below per scan line and per beam, (3) delay and apodizationvalues per dynamic range zone and per beam and (4) delayinterpolation/extrapolation coefficients per sample. The local processorcontrol C-210 also controls the sampling rate of the ADC R-118.

a. I/O Processor

With respect to the secondary or local control C-210, the controller andI/O processor C-260 controls all of the read and write operations.

b. Memory Address and Delay Processor

In a preferred embodiment, the memory address and delay processor C-264calculates an interpolated and/or extrapolated delay value for eachoutput sample of each beam of its associated beamformer processor R-120,using zone boundary delay values and the interpolation and/orextrapolation coefficients (α_(range)) which are provided by the centralcontrol C-104 through a primary delay processor of a focus controlC-132. The zone boundary delay values are defined for example by delayprofiles (FIG. 5c) at specified range boundaries. The coefficients,α_(range), allow for interpolation (and/or extrapolation) in rangebetween (and/or outbound of) the delay profile boundaries in order toincrease the density of delay values between the range boundaries. Ascan be appreciated, each digital multi-channel receive processor R-120has a memory address and delay processor C-264 associated with it inorder to afford the dynamic focusing of the invention. For multiple beamoperation, delay interpolations are time interleaved.

The delay processor C-264 performs local interpolation/extrapolation inorder to increase the density of the sparse, decimated delay profiledata set communicated to the memory address and delay processor C-264from the focus processor C-132 of the central control C-104. After theinterpolation/extrapolation step in interpolator C-199 (FIG. 4), thedelay value is divided with the most significant bits (coarse delay)being sent to the time delay memory R-152 in order to facilitate theselection of samples for desired beam or beams. The least significantbits (fine time delay) of the time delay value is sent to the phase andfrequency processor C-266 where it is turned into a phase value asdescribed more fully hereinbelow.

If selected, the architecture provides for a delay calibration valuewhich can be added to the delay data prior to interpolation ininterpolator C-199. The digital receive path delay calibration valuesfrom the calibration processor C-262 are supplied on line via C-195 tointerpolator C-199.

Alternative embodiments can have less than a one- to-one relationshipbetween beamformer processor R-120 and memory address and delayprocessor C-264 and be within the spirit of the invention. Further, suchcoefficients α_(range) can be locally generated by the memory addressand delay processor C-264. Further it is to be understood that stilldifferent delay value generation schemes can be employed and be withinthe spirit of the invention. By way of example, an accumulator structuresimilar to accumulator C-272 of the local apodization processor C-268can be used to generate appropriate delay values.

c. Phase and Frequency Processor

The phase and frequency processor C-266 (FIG. 3,8) of local or secondarycontrol C-210 generates demodulation phase values (to, for example,account for the verniering of F_(o) by the transmit beamformer system),and also phase shift correction values determined by the central controlsystem C-104. The demodulation phase values are ideally calculated as anintegration of the demodulation frequency (FIGS. 10a, 10b and 10c)generated from the frequency profile generator C-141. As hardware thataccomplishes such integration is expensive, the demodulation phasevalues are preferably calculated as the sum of (1) a product, computedin multiplier C-140 of the demodulation frequency specification profilesf_(D) (t) FIGS. 10d, 10e, and 10f, from the frequency profile generatorC-141 and a demodulation reference time from the memory address anddelay processor C-264 synchronized with the input of data to the delaymemory R-152 and (2) a constant value added by adder C-141, as morefully explained below.

The fine focusing phase correction values, as computed in multiplierC-138, are the product of the instantaneous phase correction frequencyf_(p) (t) from the frequency profile generator C-141 (FIGS. 10a, 10b and10c) and the residual or fine delay time (LSBs of delay time) from thememory address and delay processor C-264. Both the demodulationfrequency and the phase correction frequency used in computing thefocusing phase values are computed by choosing, in a preferredembodiment, one of the respective frequency profiles generated in thefrequency profile generator C-141. The two phase values, the fine phaseshift value and the demodulation phase value are added by summer C-142and communicated to a look-up table C-144 where the phase value isconverted into a complex I/Q value.

In a preferred embodiment all demodulation to or near baseband occurs inthe complex multiplier. However, in other situations such as by way ofexample only, where there are frequency offsets, such demodulation canoccur alternatively in decimator two through the use of complex filtercoefficients with residual demodulation occurring in the complexmultiplier. Such frequency offsets can, by way of example only, resultwhen the carrier frequency is verniered from the receive signal nominalcenter frequency F₀ by the above referenced digital transmit beamformersystem T-100. Such verniered center frequency can be the same for allbeams transmitted from the transmit beamformer T-100 or different foreach of multiple transmit beams.

The frequency for demodulation and for phase shift or rotation can beindependently programmed in order to select one of the following threefrequency-vs-time profiles:

(1) The frequency remains at a constant start frequency F_(start)(generally the carrier frequency F_(c)) which is time independent asshown in FIG. 10a; or

(2) The frequency is shifted down from the start frequency (F_(start))by downshift slope ΔF_(downslope), until it either: (a) saturates at aconstant limit frequency, F_(limit), in one embodiment, or (b) reaches aspecified time limit, T_(break), and thereafter remains at a constantfrequency as shown in FIG. 10b; or

(3) The frequency is first shifted down from the start frequency,F_(start), by a downshift slope, ΔF_(downslope), until it either: (a)saturates at a constant limit frequency, F_(limit), in one embodiment,or (b) reaches a specified time limit, T_(break), and thereafter isimmediately shifted up by an upshift slope, ΔF_(upslope), until thefrequency either: (a) saturates at the start frequency, F_(start),or (b)is allowed to continue without saturating at the start frequency (FIG.10c).

Both the demodulation frequency, f_(D) (t), and the frequency f_(P) (t)applied to generate the focusing phase shift value, can be selected fromany of the above similar frequency profiles. Thus, the same profile canbe applied to both multipliers C-138 and C-140. Different frequencyprofiles can also be applied to these multipliers and fall within thescope of the invention.

These profiles model frequency attenuation of ultrasound signalstransmitted through tissue. Thus, for example, the longer that abroadband signal is propagated through tissue, the more that the centerfrequency of the signal will be downshifted due to such attenuation. Inthis embodiment, all the profiles began at frequency F_(start). Thisfrequency can be the carrier frequency F_(c) of the receive beam. It isunderstood that although the transmit carrier frequency and thecorresponding receive carrier frequency can be the same, there is norequirement that they are in fact the same. Accordingly, the startfrequency of the frequency profiles can be that of the center frequencyof the receive beamformer should it be different from that of the centerfrequency of the transmit beamformer. Accordingly F_(start) can be anyvalue. However, F_(start) is preferably the transmit carrier frequencyF_(c) which is equal to the vernier factor times the center frequency,vF_(o).

The parameters for defining the above frequency profiles are stored inthe central control C-104. The frequency profile generator C-141 of thephase and frequency processor C-266 receives these parameters andcalculates the frequency values on a receive- sample-by-receive-samplebasis. These frequency values define the frequency profiles of FIGS.10a, 10b and 10c.

For one embodiment, the parameters downloaded from the central controland programmed into the local control include the start frequency, thefrequency limit, the frequency downslope, and the frequency upslope. Asindicated above, the start frequency is generally the carrier frequencyF_(c). The frequency limit is the lowest frequency value used for theabove calculations. It is understood that the numbers stored in thecentral control C-104 can be updated at any time based on new data whichcan, for example, be introduced and stored on the central control C-104for example, from hard disk memory.

In another preferred embodiment, the downloaded parameters include thestart frequency, the break time, T_(break) the frequency downslope andthe frequency upslope. In this embodiment, the downslope is limited notby a limit frequency but by time, T_(break). Thus, the frequency profilein FIG. 10c is allowed to slope down until the T_(break) has expired. Atthat point, the frequency profiles slopes up.

Preferably, the phase and frequency processor C-266 calculates allprofiles simultaneously and then the central and/or local processorcontrol selects the frequency profile, based on criteria pre-stored inthe central control C-104, for each imaging mode, to calculate ademodulation phase value and a residual time delay phase value in orderto provide the most optimally enhanced image.

Additionally, it is understood that in a multiple beam situation, eachof the beams can be received with a different carrier frequency, F_(c).The central processor could, for example, select different frequencies,slopes, and time limits for each of the beams in order to provide for anenhanced image. In such a situation, the start frequencies for each ofthe above three frequency profiles would depend upon the frequency forthe particular beam formed by the beamformer processor. Thus thefrequency profiles for each beam could be specified with entirelydifferent parameters.

As indicated above, as preferably implemented, the demodulation phasevalue is the sum of (1) a product in multiplier C-140 of thedemodulation frequency f_(D) (t) (FIGS. 10d, 10e, and 10f) from thefrequency profile generator C-141 and a demodulation reference time tfrom the memory address and delay processor C-264 and (2) a value addedby adder C-141. If the reference time t is given by 0≦t≦T_(break), thenmultiplexer C-143 causes t to be multiplied by f_(D) (t) at multiplierC-140 and multiplexer C-145 causes a zero value to be added by adderC-141. Accordingly, the demodulation phase value is f_(D) (t) t. If, onthe other hand, the reference time t is given by T_(break) ≦it thenmultiplexer C-143 causes t-T_(break) to be multiplied by f_(D) (t) andmultiplexer C-145 causes the constant value f_(D) (T_(break))·T_(break)(see discontinuities in FIGS. 10e and 10f) to be added to the result.Accordingly, the demodulator phase value is f_(D)(t)·(t-T_(break))+f_(D) (T_(break))·T_(break).

d. Apodization Processor

The apodization processor C-268 (FIG. 8) obtains a sparse table of rangebounded apodization values from the focus processor C-132 of the centralcontrol C-104. Also obtained from the central control C-104 is the zonewidth 2^(B) between the range bounded apodization value, which zonewidth is specified by a value B. If one zone boundary apodization valueis A₁ (FIG. 11) and the other zone boundary apodization value is A₂,then the accumulator C-272 (FIG. 8) of apodization processor C-268 cangenerate incremented apodization values between A₁ and A₂ by preferablyadding ##EQU1## to the accumulated apodization values (with the startingvalue being A₁). Accordingly, apodization values are generated every2^(B) intervals between A₁ and A₂ in order to fill out the sparse dataset sent by the central control. This above operation is implicitly alinear interpolation. However, nonlinear techniques can also beimplemented as well as extrapolation techniques.

Alternatively, it is to be understood that local apodization processorC-268 can internally calculate the interpolation/extrapolation rangecoefficients in a local range coefficient generator based on scangeometry parameters supplied from the central control C-104. Theseparameters define the particular scanning format that is being used.Further in still other embodiments such apodizationinterpolation/extrapolation coefficients can be pre-stored in thecentral control and downloaded to the local apodization processor.

The apodization processor C-268 calculates an interpolated/extrapolatedapodization value for each output sample of each beam. To supportmultiple beam operation, the apodization processor C-268 interleavesinterpolation/extrapolation calculations. As with the delay values, theapodization values, if desired, can be modified by supplying theapodization calibration values from the calibration processor before theapodization value is applied to the complex multiplier.

The complex value representation of the phase shift and the apodizationvalues, multiplied together in multiplier R-260, are sent to the complexmultiplier R-156 to be multiplied with the complex sample signal value.

e. Calibration Processor

The calibration processor C-262 is activated when a scan format ortransducer is changed. During calibration, a common calibration signalfrom, for example, the transmit beamformer system T-100 is injected intoall receive channels. The component tolerances in analog circuitry priorto digitization in ADC R-118 can result in analog-path-to-analog-pathsignal variances. The local calibration processor compares the outputsignal to a fixed calibration reference value which is stored in thelocal calibration processor. The local calibration processor computesdelay and apodization correction values for the local control in orderto drive the difference between the output signals and the referencesignal to zero through an iterative process.

These correction values are sampled on an analog signal path basis andsupplied by the calibration processor C-262, with respect to magnitude,to the apodization processor C-268 and, with respect to delay and phase,to the memory address and delay processor C-264.

For operations including, by way of example only, sliding aperture,random aperture and synthetic aperture, multiple fixed calibrationreference values can be stored.

In addition to the above locally computed calibration values,calibration values can be downloaded from the central control. Forexample, calibration values for each type of transducer can bepre-stored in central control or provided to central control when a newtransducer is selected. Such calibration values can then be downloadedto the local calibration processor to be combined with the locallygenerated calibration values, if appropriate.

4. Final Beamformation Processor (Baseband Multi-Beam Processor)

The digital multi-channel receive processors R-120 are summed by summerR-126 and the results communicated to the baseband multi-beam processorR-125 (FIG. 2b) which comprises a baseband filter and phase alignerR-127 (FIGS. 2b, 3, 9) and a baseband processor control C-270 (FIGS. 2b,3, 9).

a. Subarray Summer

As is known in the art, two standard methods to sum multiple inputs areparallel summation and sequential summation. The present embodimentcombines aspects of these two approaches for a fast and efficientsummation process. FIG. 2b depicts the summation process of the presentdigital receive beamformer system. Pairs of digital multi-channelreceive processors R-120 are combined through parallel summation. Eightpairs of processors R-120 are sequentially summed by a subarray summer(subarray sums block R-126, FIG. 2b). The summers for this firstsummation step can be accomplished outside the processors R-120.Alternatively, the processors R-120 can include summers to effect thisstep.

After the above summation, then four such subarray sums are summed inparallel by a semi-final summer (final sums block R-126). Following thisstep is a parallel summation step where the sums from two semi-finalsummers are summed in parallel in final summer (sum block R-126). It isto be understood that alternative combinations of parallel andsequential summation techniques or all parallel summation or allsequential summation techniques could be used and be within the scope ofthis invention.

b. Baseband Filter and Phase Aligner

The complex baseband signal (or signals in the multiple beam case) fromthe digital multi-channel receive processors R-120 which represent thesummation of all the signals from the elements sampled across the faceof the transducer, is communicated to a baseband filter and phasealigner block R-127. Block R-127 includes a baseband filter R-250 (FIG.9) which performs filtering and rational sample rate conversion(interpolation and decimation). Block R-127 also includes a phasealigner R-252 (FIG. 9) which provides for (1) scan-line-dependent andrange-dependent phase adjustments of the signal required to correct forphase differences resulting from line-to-line apodization changes, scangeometry, and nonaligned effective transmit and receive origins, (2)remodulation (frequency alignment) of the signal to correct for phasedifferences resulting from different transmit frequencies per scan line,and (3) gain adjustment per scan line. The advantage of the use of ascan-line-to-scan-line variable frequency mode on transmit and receivebeamformation is the reduction of grating lobes (see co-pendingapplication entitled: METHOD AND APPARATUS FOR ADJUSTABLE FREQUENCYSCANNING IN ULTRASOUND IMAGING, which discusses a scan-line-to-scan-linevariable frequency mode).

Such phase alignment and remodulation between desired scan lines andparticularly two or more adjacent scan lines is, for example, forpurposes of implementing coherent image processing techniques asdescribed in the above co-pending application entitled: METHOD ANDAPPARATUS FOR COHERENT IMAGE FORMATION.

Thus, the purpose of the phase aligner is to maintainscan-line-to-scan-line coherency for (1) adjustable frequency operation,(2) synthetic scan line operation, as well as for (3) synthetic apertureoperation, and (4) future operations on coherent beam samples.

Baseband filter R-250 preferably includes a multi-tap FIR filter whichis programmable with both real and complex coefficients h4, and arational sample rate converter. The rational sample rate converterincludes an interpolator which has an integer upsampling factor L and adecimator with an integer down sampling factor M. Baseband filter R-250accordingly accomplishes the following tasks.

First, baseband filter R-250 increases the signal-to-noise ratio byrejecting out-of-band noise frequencies, and/or maximizing thesignal-to-noise ratio with a matched filter or quasi-matched filterdesign, preferably for matching to substantially Gaussian transmitpulses as well as pulses of other shapes. Gaussian pulses are especiallyuseful as they represent waveforms that do not distort duringtransmission through attenuative media such as the body.

Second, baseband filter R-250 enables pulse equalization and shaping bycompensating for both the transducer frequency response and the analogsignal path prior to the ADC R-118.

Third, baseband filter R-250 performs a sample rate conversion(decimation function) based upon the rational (non-integer) decimationfactor L/M (where L and M are integers). Accordingly, the sample rate isconverted to a rate that is advantageous for an image display.

Examples of such decimation can be found in the references identifiedwith the above discussion of decimator one and decimator two. The filtercoefficients and non-integer decimation factors for baseband filterR-250 are programmed into baseband filter/phase aligner R-127 by beingdownloaded from the central control C-104 to coefficient and rate memoryC-278. The downloaded coefficients and factors can be changed at anytime by introducing new coefficients and factors into the centralcontrol C-104. The coefficients and factors stored in the coefficientand rate memory C-278 are selectable by the central control C-104 forprogramming the filter and decimation ratio L/M of the baseband filterR-250.

The complex multiplier R-254 of phase aligner R-252 operates in a mannersimilar to complex multiplier R-156 (FIG. 3).

Following complex multiplier R-254 is a register C-296 which stores scanline sample data so that it can be reported to the DMA processor C-202of the central control C-104 for providing scan-line-to-scan-linecalibration.

c. Baseband Processor Control

The phase aligner includes a control function which is contained in abaseband processor control C-270 (FIGS. 2b, 3, 9). In this basebandprocessor control C-270, a scan-line-to-scan-line or beam-to-beam gainadjustment value and a phase adjustment value are generated in a timeinterleaved manner. As discussed above, the phase correction value isthe sum of the phase terms including: (1) a phase adjustment termrequired to correct for phase differences due to scan-line-to-scan-lineapodization changes, and scan geometry which results in non-alignedeffective transmit and receive origins (the scan-line-dependent andrange-dependent phase adjustment term) and (2) a phase term required toremodulate the signal as though each line had used a common carrierfrequency. As discussed herein and in copending U.S. patent applicationsentitled: METHOD AND APPARATUS FOR TRANSMIT BEAMFORMER SYSTEM and METHODAND APPARATUS FOR ADJUSTABLE FREQUENCY SCANNING IN ULTRASOUND IMAGING,using a frequency scaling factor or frequency vernier factor, each beamcan have a different carrier frequency. The phase aligner accordinglyprovides for remodulation between beams so that all beams are adjustedfor differences in carrier frequencies.

In operation a source data set including scan format geometryparameters, sparse scan line gain and delay value, interpolationcoefficient and non-integer decimation factors are downloaded from thecentral control C-104 to the baseband processor control C-270.Additionally, frequency parameters used in the frequency profilegenerator of the central control C-104 in accordance with FIGS. 10a, 10band 10c are downloaded to the baseband processor control C-270.

The baseband processor control C-270 of FIG. 9 includes a gain and phaseRAM C-280, a line interpolator C-282 which is supplied withpre-calculated and pre-stored line interpolation coefficients (α_(line))by the central control C-104, and a range interpolator C-284 with arange accumulator C-286, which is supplied with a rational decimationfactor L/M and a phase zone width, both of which values arepre-calculated and pre-stored in the central control C-104. The rationaldecimation factor L/M is the same value supplied to the baseband filterR-250. Accumulator C-286 operates in the same manner as does accumulatorC-272 of the local apodization processor C-268 (FIG. 8). Additionally asis known in the art a sample rate conversion in accordance with therational decimation factor L/M is accomplished in order to match thesample data rate of the baseband filter R-250.

Alternatively the range interpolator/extrapolator C-284 can be suppliedwith programmable (as defined above) interpolation/extrapolationcoefficients which are, by way of example, either (1) pre-calculated andpre-stored in or calculated by the central control or (2) calculatedlocally in baseband processor control C-270 by a coefficient generator.

The baseband processor control C-270 also includes a remodulationfrequency processor C-292 which is preferably implemented as a doublephase accumulator. The double phase accumulator calculates phaseadjustment values to correct for line-to-line frequency differences andthus to remodulate the signal as though a common carrier frequency hadbeen used across all scan lines.

From the central control C-104, pre-calculated and pre-stored valuesrepresenting the frequency differences between scan lines (deltafrequency values) are sent to the remodulation frequency processorC-292. These frequency difference values are based on frequencies andfrequency slopes such as specified in FIGS. 10a, 10b and 10c. By way ofexample only, let it be assumed that the frequency profiles for two scanlines look like FIG. 10b but with different start frequency, F_(start),values and different downshift slope, ΔF_(downslope), values.Accordingly, downloaded to baseband processor control C-270 from thecentral control for the two scan lines are the difference in frequenciesbetween the scan lines and the difference in the rate of change of thefrequency profiles over time. These values are calculated by theacquisition processor C-130 based on stored parameters and dependentupon the particular rational conversion factor L/M currently being used.The first accumulator of processor C-292 accumulates the difference inthe rates of change of the frequency profiles over time between scanline while the second accumulator accumulates the difference in thefrequencies between the scan lines over time. If there is no differencein the rate of change of the frequency profile over time, (i.e. theprofile are the same exact for initially different F_(start) values, orafter T_(break) in FIG. 10b when the slope goes to zero) the firstaccumulator performs no function. With no difference in the rate changesof the frequencies between the scan lines, only the second accumulatoraccumulates the frequency differences over time resulting in acorrective remodulation phase value.

The phase adjustment due to scan-line-to-scan-line apodization changes,scan geometry which results in non-aligned transmit and receive origins,and the phase adjustment due to remodulating the signal to an effectivecommon carrier frequency are added in a summer C-288 and the summedphase value is then converted in a look-up table C-290 to sine andcosine representations. As part of the look-up table C-290 function, thegain is multiplied by the sine and cosine representations. This value isapplied to complex multiplier R-252.

It is to be understood that other embodiments of the baseband processorcontrol are possible within the scope of this invention.

As indicated above the phase aligner R-127 ensures that coherent signaland sample relationships are maintained between scan lines. The transmitsamples and the echo or receive samples of the signals from beams aredefined as being coherent when sufficient information is stored,preserved, or maintained to enable the samples of the return signals tobe phase and amplitude corrected from scan-line-to-scan-line. Theprocess of actually making the phase and amplitude corrections need nothave yet taken place, as long as sufficient information with respect toa reference is maintained.

When a signal sample is processed coherently, the processing continuesto maintain sufficient information to perform phase and amplitudecorrection at a later time. When two or more samples are processedcoherently (e.g., coherently summed), the phase and amplitudecorrections necessary for phase and amplitude alignment must havepreviously been performed.

Coherent processing of two or more signal samples yields significantbenefits, such as being able to calculate synthetic samples, asdescribed in the above co-pending application.

Due to the beamformer control C-104 specifying and accounting for allaspects of the transmit and receive signal, the entire system maintainsall signal samples as coherent samples throughout the transmit andreceive signal path, until the signal is finally detected in anoperation which is external to beamformation.

It is to be understood that although scan-line-to-scan-line phasealignment is accomplished by baseband filter/phase aligner R-127 afterbeamformation, that such phase alignment can be provided prior tobeamformation in the digital multi-channel receive processor R-120. Byway of example, such phase alignment can be accomplished in each complexmultiplier R-156 of each processor R-120.

5. Synthetic Aperture

Synthetic aperture, in the preferred embodiment of this invention, ischaracterized by: (1) partitioning the array of transducer elements intoa plurality of independent or substantially independent subarrays fortransmission and/or reception, each subarray consisting of multipletransducer elements; (2) executing a plurality of transmit/receivesequences with a subarray pair; (3) for each sequence, acquiring thecoherent samples; and (4) combining, preferably by summation or weightedsummation, all corresponding coherent samples. With such an arrangement,the number of transmit and/or receive electronic paths is effectivelyincreased, and the transducer aperture on transmission and/or receptionis increased.

Synthetic aperture scanning is described, for example, in Klahr U.S.Pat. No. 3,805,596, entitled: "High Resolution Ultrasonic ImagingScanner," and in Saugeon U.S. Pat. No. 4,733,562, entitled: "Method AndApparatus For Ultrasonic Scanning Of An Object." Synthetic aperturescanning is also identified in Kino, "Acoustic Imaging forNondestructive Evaluation," and Sutton, "Underwater Acoustic Imaging,"both in Proceedings of the IEEE, Vol. 67, April 1979. All the abovereferences are incorporated herein by reference.

The present digital receive beamformer system supports syntheticaperture operations. The calibration processor stores gain and phasecorrections for the subarray receive apertures used in the syntheticaperture operations. The phase aligner maintains coherency of theseparately received subarray receive aperture beams so that a summer cancombine the signals to form a synthetic aperture scan line.

Further description of such synthetic aperture operation can be found inco-pending application entitled: METHOD AND APPARATUS FOR COHERENT IMAGEFORMATION.

The present invention provides significant enhancements over the priorart by providing for dynamic focal-point-to-focal-point time delayfocusing and dynamic apodization of the image scanned across theaperture of the transducer and along the range direction.

This system further provides for multiple beam capacity with a singlecomputational path for each of the digital multi-channel receiveprocessors assigned to one or more transducer elements. The multiplebeams superimposed in the data are not distinguished until late in thedigital signal processing path affording further computationalefficiencies. The present invention provides for trade-offs notavailable in the prior art allowing the full computational bandwidth ofeach digital signal path to be utilized. The trade-offs afforded arethose among center frequency, spatial range resolution, and the numberof beams selected.

The digital multi-channel receive processor R-120 has four main signalpath functional components which are arranged in a novel order for amongother reasons the above specified computational efficiencies. Theprocessor R-120 includes first the decimator one, second the memory,third the decimator two, and fourth the complex multiplier which operatein the novel first through fourth order presented.

Due to this flexibility in the computational bandwidth, the presentsystem is highly programmable and thus a wide variety of imagingapplications and a wide variety of imaging transducers can be used withthe system. For example, if high resolution is desired, then the systemcan be set for 10MHz operation with one beam in operation (Table 1) anda 10MHz transducer can be attached to the system. If, for example, ahigher image frame rate is desired so that movement of an image, such asa heart valve, in real time can be observed, then the system can be setfor, by way of example, four beams, each with a per beam centerfrequency of 2.5MHz. Such a higher frame rate would be beneficial forcolor Doppler imaging (F-mode) of moving body fluid such as blood. Insuch application resolution would not be as important as the ability torepresent the fluid flow.

The present invention also includes a local processor control which candynamically generate delay, phase apodization and calibration correctionvalues for enhanced beamformer performance.

The invention further includes a phase aligner for correcting phasepost-beamformation on a beam-to-beam basis. Further, the entire systemensures that the samples are maintained in a coherent manner.

While the present embodiment has been described with respect toreflective or echo receive signals, the present system could as well beoperable with transmission imaging having receive signals that passthrough the object to be imaged.

Other objects, aspects and advantages of the invention can beascertained from the figures and claims appended.

It is to be understood that other embodiments of the present inventioncan be fabricated and fall within the spirit and scope of the invention.

We claim:
 1. An ultrasound receive beamformer comprising:an adjustableresolution digital processor for processing a signal which is used toconstruct N receive beams, where N is a variable positive integer; saiddigital processor comprising means for adjusting resolution of saidsignal as a function of N in order to utilize overall digital processingcapability of the digital processor.
 2. The beamformer of claim 1whereinsaid adjusting means comprises means to adjust spatial range resolutionas a function of N.
 3. The beamformer of claim 2 whereinsaid adjustingmeans comprises means to adjust spatial range resolution relative to anumber of concurrent receive beams per transmit event and the nominalcenter frequency of each receive beam.
 4. The beamformer of claim 1inwhich said digital processor includes a decimator for downsampling saidsignal depending on a selected resolution.
 5. The beamformer of claim1in which said adjusting means is operative to adjust the resolution ofsaid signal dependent additionally on a nominal center frequency of thesignal.
 6. The beamformer of claim 1 wherein:said digital processorprocesses samples of said signal to construct components of at least onereceive beam in a single processor, associating each beam with a channelin said processor.
 7. The beamformer of claim 1 wherein:said digitalprocessor includes a multiplicity of digital multi-channel receiveprocessors with each digital multi-channel receive processor processingsamples of said signal to construct components of at least one receivebeam, associating each receive beam with a channel in said digitalmulti-channel receive processor.
 8. The beamformer of claim 1whereinsaid adjusting means comprises means to adjust the resolution of saidsignal depending on the number N of beams which are formed from saidsignal in order to utilize full performance of the processor.
 9. Anultrasound receive beamformer comprising:a variable resolution digitalprocessor for processing a signal which is used to construct at leastone receive beam, said signal characterized by a variable nominal centerfrequency; said digital processor comprising means for adjustingresolution of the signal as a function of the nominal center frequencyin order to utilize overall digital processing capability of the digitalprocessor.
 10. The beamformer of claim 9 wherein:each of said receivebeams is associated with a carrier frequency derived from the commonnominal center frequency; and said adjusting means is operative toadjust the resolution of said signal depending on the nominal centerfrequency.
 11. The beamformer of claim 9 wherein:said adjusting means isoperative to process samples of said signal to construct components ofat least one receive beam in a processor, associating each beam with achannel in said processor.
 12. The beamformer of claim 9 wherein:saiddigital processor includes a multiplicity of digital multi-channelreceive processors with each digital multi-channel receive processorprocessing samples of said signal to construct components of at leastone receive beam, associating each beam with a channel in saidprocessor.
 13. An ultrasound receive beamformer for receiving a signalfrom a transducer array which is used to construct at least two beams,said beamformer comprising:a memory for storing a first set of datasamples representative of the signal received during a first timeinterval from the transducer array, which stored data samples make nodistinction between the at least two beams; said memory including meansfor applying a different set of time delay values associated with eachbeam to the first set of data samples stored in the memory to providereconstructed data samples which distinguish between the at least twobeams.
 14. The beamformer of claim 13 whereinsaid memory outputs thereconstructed data samples in an interleaved manner.
 15. The beamformerof claim 13 further comprising:a decimator for further processing thereconstructed data samples of the beams in an interleaved manner. 16.The beamformer of claim 13 further comprising:a decimator for furtherprocessing the reconstructed data samples of the beams in a timemultiplexed manner such that overall computational capability of thedecimator is utilized; and further in which said decimator downsamplesthe reconstructed data samples depending on at least one of a number ofbeams being processed, spatial range resolution desired for each beam,and a nominal center frequency.
 17. The beamformer of claim 13 furthercomprising:a decimator for downsampling the reconstructed data samples;and a phase rotator for applying a phase rotation to the reconstructeddata samples.
 18. The beamformer of claim 17 including:a unit forapodizing the reconstructed data samples.
 19. The beamformer of claim 13including:a decimator for further processing the reconstructed datasamples of the beams such that overall computational capacity of saidbeamformer is utilized; and further in which said decimator downsamplesthe reconstructed data samples depending on at least one of a number ofbeams being processed, spatial range resolution desired for each beam,and a nominal center frequency.
 20. A method for processing a receivedand digitized signal representative of a waveform sensed by elements ofan ultrasound transducer array comprising the steps of:processingseparately the received and digitized signal representative of thewaveform sensed by each element of the ultrasound transducer array; andadjusting processed resolution of the received and digitized signal inresponse to a variable indicative of a number of beams which are formedfrom the received and digitized signal in order to utilize overallprocessing capability of the processing step.
 21. The method of claim20in which said adjusting step further comprises the step of adjusting aspatial range resolution of the received and digitized signal dependingon a number of beams which are formed from the received and digitizedsignal.
 22. The method of claim 20in which said adjusting step furthercomprises downsampling the digital signal depending on a number of beamswhich are formed from the digitized signal.
 23. A method for processinga received and sampled signal representative of a waveform sensed byelements of an ultrasound transducer array comprising the stepsof:processing separately the received and sampled signal representativeof the waveform sensed by each element of the ultrasound transducerarray; and adjusting processed resolution of the received and sampledsignal in response to a variable indicative of a nominal centerfrequency of the received and sampled signal in order to utilize overallprocessing capability associated with the processing step.
 24. Themethod of claim 23 wherein said signal is used to form at least onebeam, and further wherein said beam is associated with a carrierfrequency derived from a common nominal center frequency, the methodfurther comprising:adjusting the resolution of the received and sampledsignal depending upon the nominal center frequency.
 25. The method ofclaim 23 wherein the adjusting step includes:adjusting a spatial rangeresolution of the received and sampled signal.
 26. A method forprocessing a received and digitized signal representative of a waveformwhich is used to construct at least two beams sensed by elements of anultrasound transducer array comprising the steps of:storing a first setof data samples representative of the signal received during a firsttime interval from the ultrasound transducer array, which stored firstset of data samples makes no distinction between the at least two beamsbeing constructed; applying a different set of time delay valuesassociated with each beam to the stored first set of data samples toprovide reconstructed data samples which distinguish between the atleast two beams being constructed.
 27. The method of claim 26 furthercomprising:processing the reconstructed data samples of the beams beingconstructed such that an overall computational capability for suchprocessing is utilized; and downsampling the reconstructed data samplesdepending on a number of beams being processed and a resolution desiredfor the processed signal.
 28. The method of claim 26 furthercomprisingthe step of performing at least one of filtering anddownsampling the reconstructed data samples; and applying a phaserotation to the reconstructed data samples.
 29. The method of claim 28whereinsaid step of downsampling the reconstructed data samples isfurther characterized as downsampling the signal depending on a numberof beams represented by the reconstructed signal and a resolutiondesired for the processed signal.
 30. The method of claim 28including:apodizing the reconstructed data samples.
 31. The method ofclaim 26 in which processing of the reconstructed data samples of thebeams is performed in manner such that an overall computational capacityis utilized; anddownsampling the reconstructed data samples dependingupon a resolution desired for the processed signal.
 32. The method ofclaim 26 further comprisingthe step of selectively downsampling thereconstructed data samples depending upon on resolution desired for theprocessed signal.
 33. The method of claim 26 furthercomprising:downsampling the reconstructed data samples depending on anumber of beams being processed.